Low inductance power converter with compact commutation cell

ABSTRACT

A power electronics converter may include a converter commutation cell having a power circuit and a gate driver circuit. The power circuit includes at least one power semiconductor switching element and at least one capacitor. Each power semiconductor switching element is included in a power semiconductor prepackage, each prepackage including one or more power semiconductor switching elements embedded in a solid insulating material, each power semiconductor switching element having at least three terminals including a gate terminal. The gate driver circuit is electrically connected to and configured to provide switching signals to the gate terminal of each of the at least one power semiconductor switching element. A peak rated power output of the power electronics converter is greater than 25 kW and a value of a converter parameter α is less than or equal to 5 pHm3.

The present patent document is a continuation patent application of U.S.patent application Ser. No. 17/899,960, filed Aug. 31, 2022, whichclaims the benefit of German Patent Application No. DE 10 2022 205483.0, filed May 31, 2022, which are hereby incorporated by reference intheir entirety.

TECHNICAL FIELD

This disclosure relates to power electronics converters, andparticularly but not exclusively to power electronics convertersincluding power semiconductor prepackages for use in the power andpropulsion systems of aircraft.

BACKGROUND

In aerospace, aircraft and power and propulsion systems of the aircraftare becoming increasingly electrical in their design. Some proposedplatforms are purely electric, relying completely on batteries or fuelcells for all their power and propulsion requirements. Other proposedplatforms are of the hybrid-electric type, and others still are ‘moreelectric’ in that the proposed platforms derive most or all theirpropulsive power from on-board engines (e.g., gas turbine engines) buthave an increased number of electrically powered aircraft and enginesystems, sub-systems, and accessories.

The electrical power systems that feature in these platforms includepower electronics converters. AC-DC converters (e.g., inverters andrectifiers) convert between AC and DC, for example, to deliver AC to anelectrical machine configured as a motor from a DC power source (e.g., abattery or DC power channel), or to deliver DC power from an electricalmachine configured as a generator to a DC power channel or rechargeablebattery. DC-DC converters may be used to regulate the DC voltagedelivered from a battery to a DC power channel, for example. Theelectrical power systems may also include other power electronicsdevices such as, for example, protection devices such as solid-statepower controllers (SSPCs) and solid-state circuit breakers, some ofwhich may be incorporated into the converters themselves.

So-called power modules, or power electronics modules, are the dominantstate of the art technology for power electronics converters. In aconverter including a power module, the components of the convertercircuit, which include power semiconductor devices including transistorsand diodes, and smoothing DC-link or input capacitor(s), are fixed to acarrier substrate and electrically connected to each other (e.g., usingwire bonds). Power module-based converters are widely used in, forexample, the automotive industry and are used in existing aerospacesystems, where the power module-based converters may provide acceptableperformance with average operating efficiencies as high as 95%.

The performance of converters based on the power module topology isacceptable for most applications, especially when due account is takenof their relatively low cost and high availability. From the point ofview of aerospace applications, however, improving the efficiency andpower-to-weight ratio of power electronics converters would beadvantageous. Compared with ground-based applications includingautomotive applications, aerospace applications are highly sensitive toweight. For purely electric aircraft applications especially, even arelatively small increase in converter efficiency could providemeaningful improvements in aircraft performance and mission range.

Improvements in the performance of converters based on the existingpower module topologies are likely to be limited to what can be achievedthrough advances in the underlying semiconductor technologies. This isat least in part because of the inherently high parasitic inductance ofthe commutation cell of a power module, much of which is introduced bythe electrical connections between the commutation cell components.Parasitic inductance in the commutation cell is associated withtransistor switching losses and voltage overshoot during turn-off, whichnot only limits efficiency and generates heat that is to be removed, butalso limits other performance characteristics including the transistorswitching frequency.

SUMMARY

The scope of the present disclosure is defined solely by the appendedclaims and is not affected to any degree by the statements within thissummary. The present embodiments may obviate one or more of thedrawbacks or limitations in the related art.

A power electronics converter is disclosed. The power electronicsconverter includes a converter commutation cell including a powercircuit and a gate driver circuit. The power circuit includes at leastone power semiconductor switching element and at least one capacitor.Each power semiconductor switching element has at least three terminalsincluding a gate terminal. The gate driver circuit is electricallyconnected to and configured to provide switching signals to the gateterminal of each of the at least one power semiconductor switchingelements. The power electronics converter may be an AC-DC powerelectronics converter (e.g., an inverter or a rectifier) or a DC-DCpower electronics converter.

An electrical power system including an electrical machine and an AC-DCpower electronics converter is also disclosed. The electrical machineincludes one or more windings. The AC-DC power electronics converterincludes a commutation cell including a power circuit and a gate drivercircuit. The power circuit includes a plurality of power semiconductorswitching elements and at least one capacitor. An AC-side connection ofthe power circuit is connected to the one or more windings of theelectrical machine to supply current to or receive current from theelectrical machine. The electrical machine may be a motor or agenerator, and the AC-DC power electronics converter may be an inverteror a rectifier. The electrical machine may be a motor-generator, and theAC-DC converter may be a bi-directional converter operable as aninverter or a rectifier.

The following metric prefixes are used herein to abbreviate numericalvalues:

TABLE 1 Prefix Abbreviation Value exa E ×10¹⁸ peta P ×10¹⁵ tera T ×10¹²giga G ×10⁹ mega M ×10⁶ kilo k ×10³ centi c ×10⁻² milli m ×10⁻³ micro μ×10⁻⁶ nano n ×10⁻⁹ pico p ×10⁻¹² femto f ×10⁻¹⁵ atto a ×10⁻¹⁸

Any of the following features below may be applied singularly or incombination with each other and with the power electronics converter andthe electrical power system set out above.

Each power semiconductor switching element may be a transistor. Eachtransistor may be a MOSFET having at least the gate terminal, a sourceterminal, and a drain terminal. The MOSFETs may be Silicon Carbide (SiC)MOSFETs. In other examples, the MOSFETS are Gallium Nitride (GaN)MOSFETs.

Each power semiconductor switching element may be included in a powersemiconductor prepackage. Each power semiconductor prepackage includesone or more power semiconductor switching elements embedded in a solidinsulating material. Each power semiconductor prepackage may includeprecisely one power semiconductor switching element.

The power electronics converter may include one or more powersemiconductor logical switches each including one or moreparallel-connected power semiconductor switching elements. Each powersemiconductor logical switch may include one or more power semiconductorprepackages, each power semiconductor prepackage including at least one(and optionally precisely one) power semiconductor switching element.The number of power semiconductor switching elements per logical switchmay be greater than or equal to three. The number may be in the range ofthree to twelve.

A peak rated power output of the power electronics converter may begreater than 10 kW (equivalently kVA) and may be greater than 25 kW. Thepeak rated power may be greater than or equal to 40 kW, or greater thanor equal to 50 kW. The peak rated power may be less than or equal to 500kW. The peak rated power output may be less than or equal to 400 kW, orless than or equal to 300 kW. The peak rated power may be in the range50 kW to 300 kW.

A maximum efficiency of the power electronics converter may be greaterthan 97%. The maximum efficiency may be greater than 97.5%, greater than98%, greater than 98.5% or even greater than 99%. The maximum efficiencyof the power electronics converter may be less than 100%.

A value of a converter parameter α may be less than or equal to 5 pHm³.α is a product of the smallest cuboidal volume that encloses thecommutation cell and a parasitic inductance of the power circuit of thecommutation cell. α may be greater than or equal to 0.3 pHm³. α may beis less than or equal to 4 pHm³. α may satisfy 0.4 pHm³≤α≤3.5 pHm³. αmay satisfy 0.5 pHm³≤α≤2.5 pHm³.

The value of α divided by the peak rated power output of the powerelectronics converter may be greater than or equal to 1.5 aHm³/W. Thevalue of α divided by the peak rated power output of the powerelectronics converter may be less than or equal to 100 aHm³/W. The valueof a divided by the peak rated power output of the power electronics maybe in the range 2.5 aHm³/W to 50 aHm³/W.

A product of the parasitic inductance of the power circuit of thecommutation cell and the peak rated power output may be in the range0.05 mHW to 1.5 mHW. The product may be in the range 0.1 mHW to 1.2 mHW,or in the range 0.2 mHW to 1.0 mHW.

A value of a parameter β may be greater than or equal to 0.3 PV/s². β isa product of a maximum switching frequency of the switching signals anda maximum rate of change of a source-drain voltage of the plurality ofpower semiconductor switching elements during operation. The value of βmay be less than or equal to 10 PV/s². The value of β may be greaterthan or equal to 0.5 PV/s². The value of β may satisfy 0.8 PV/s²≤β≤5PV/s² or may satisfy 1.0 PV/s²≤β≤2.5 PV/s².

A value of a converter parameter γ may be less than or equal to 150fFs/W. γ is a total rated capacitance of the at least one capacitor ofthe power circuit divided by a product of the peak rated power output ofthe power electronics converter and a maximum switching frequency of theswitching signals. The value of γ may be greater than or equal to 1.0fFs/W. The value of γ may be less than or equal to 100 fFs/W, less thanor equal to 75 fFs/W or less than or equal to 50 fFs/W. The value of γmay satisfy 2.0 fFs/W≤γ≤50 fFs/W or may satisfy 4.0 fFs/W≤γ≤25 fFs/W.

A value of a converter parameter δ may be greater than or equal to 0.5PV/FH. δ is a maximum blocking voltage of the one or more powersemiconductor switching elements of the power circuit divided by aproduct of a parasitic inductance of the power circuit of thecommutation cell and a total rated capacitance of the at least onecapacitor of the power circuit. The value of δ may be less than or equalto 40 PV/FH. The value of δ may be greater than or equal to 1.5 PV/FH.The value of δ may be in the range 2.5 PV/FH to 25 PV/FH. The value of δmay be in the range 4.0 PV/FH to 15 PV/FH.

A value of a converter parameter ε may be greater than or equal to 10²⁶V/s⁴. ε is equal to:

$\begin{matrix}{\varepsilon = \frac{f_{\max} \times {❘\frac{dv}{dt}❘}\max}{L \times C}} & (1)\end{matrix}$

In this equation, f_(max) is a maximum switching frequency of theswitching signals, |dv/dt|_(max) is a maximum rate of change of asource-drain voltage of the one or more power semiconductor switchingelements during operation, L is a parasitic inductance of the powercircuit of the commutation cell, and C is a total rated capacitance ofthe at least one capacitor of the power circuit. The value of c may beless than or equal to 10²⁹ V/s⁴. The value of ε may be greater than orequal to 5×10²⁶ V/s⁴. The value of ε may be in the range 10²⁷ V/s⁴ to5×10²⁸ V/s⁴ or may be in the range 1.5×10²⁷ V/s⁴ to 3×10²⁸ V/s⁴.

The parasitic inductance of the power circuit of the commutation cellmay be less than or equal to 16 nH, less than or equal to 10 nH, lessthan or equal to 8 nH, less than or equal to 6 nH, less than or equal to4 nH, less than or equal to 3 nH, or even less than or equal to 2 nH.The parasitic inductance may be in the range 2 nH to 8 nH.

The total rated capacitance of the power circuit of the commutation celldivided by the peak rated power may be less than or equal to 5 nF/W, andmay be less than or equal to 3 nF/W. The total rated capacitance of thepower circuit of the commutation cell divided by the peak rated powermay be in the range 0.1 nF/W to 2.5 nF/W.

The smallest cuboidal volume that encloses the commutation cell may beless than or equal to 1,000 cm³. The smallest cuboidal volume may beless than or equal to 900 cm³, less than or equal to 800 cm³, less thanor equal to 700 cm³, or less than or equal to 600 cm³. The smallestcuboidal volume may be in the range 100 cm³ to 800 cm³, 100 cm³ to 700cm³, 100 cm³ to 600 cm³, 150 cm³ to 600 cm³, or 200 cm³ to 450 cm³.

A maximum rate of change of a source-drain voltage of the one or morepower semiconductor switching elements during operation may be greaterthan or equal to 10 kV/μs, greater than or equal to 15 kV/μs, or greaterthan or equal to 20 kV/μs. The maximum rate of change may be less than150 kV/μs, less than 120 kV/μs, less than 100 kV/μs, less than 90 kV/μs,or less than 80 kV/μs. The maximum rate of change may be in the range 10kV/μs to 120 kV/μs, in the range 15 kV/μs to 100 kV/μs, in the range 20kV/μs to 90 kV/μs, or in the range 25 kV/μs to 80 kV/μs. The maximumrate of change may be in the range 10 kV/μs to 60 kV/μs, in the range 15kV/μs to 50 kV/μs, in the range 20 kV/μs to 50 kV/μs, in the range 25kV/μs to 50 kV/μs, or in the range 30 kV/μs to 50 kV/μs. The maximumrate of change may be in the range 30 kV/μs to 40 kV/μs.

The maximum switching frequency of the switching signals (f_(max)) maybe greater than or equal to 10 kHz. The maximum switching frequency maybe greater than or equal to 20 kHz, greater than or equal to 30 kHz,greater than or equal to 40 kHz, or greater than or equal to 50 kHz. Themaximum switching frequency may be less than 100 kHz. The maximumswitching frequency may be in the range 30 kHz to 70 KHz.

The blocking voltage (e.g., the ‘source-drain blocking voltage’,sometimes referred to as the ‘rated voltage’) of each powersemiconductor switching element may be greater than 600 V, greater than700 V, or greater than 800 V. The blocking voltage may be less than1,800 V or less than 1,700V. The blocking voltage may be in the range800 V to 1,600 V, 900 V to 1,500, or 1,000 V to 1,400 V.

The power electronics converter may further include a multi-layer planarcarrier substrate. The multi-layer planar carrier substrate may definean x-y direction parallel to a planar surface of the substrate and az-direction perpendicular to the x-y direction. The carrier substratemay include a plurality of electrically conductive layers extending inthe x-y direction and at least one electrical connection extending inthe z-direction. The carrier substrate may include an outer conductivelayer on one or both of its opposed planar surfaces.

The multi-layer planar substrate may be a rigid printed circuit board(PCB). The multi-layer planar substrate may be a flexible PCB. Themulti-layer planar substrate may be a ceramic carrier substrate. Themulti-layer planar substrate may be a structural component of theconverter.

Each power semiconductor prepackage may further include at least oneelectrical connection extending in the z-direction from at least oneterminal of each of the one or more power semiconductor switchingelements through the solid insulating material to an electricalconnection side of the power semiconductor prepackage. At least one ofthe terminals of each of the one or more power semiconductor switchingelements of the prepackage may be connected to at least one of theconductive layers of the multi-layer planar carrier substrate at theelectrical connection side of the power semiconductor prepackage.

The electrical connection side of the power semiconductor prepackage maybe spaced apart in the z direction from the multi-layer planar carriersubstrate so as to define a gap (referred to herein as the prepackagegap) between the multi-layer planar carrier substrate and the electricalconnection side of the prepackage. A size in the z-direction of theprepackage gap may be less than or equal to 300 μm. The size of theprepackage gap may be less than or equal to 250 μm. The size of theprepackage gap may be less than or equal to 200 μm. The size of theprepackage gap may be less than or equal to 150 μm. The size of theprepackage gap may be greater than or equal to 10 μm, or greater than orequal to 20 μm, or greater than or equal to 50 μm, or greater than orequal to 80 μm. The prepackage gap may be in the range 20 μm to 250 μm,or in the range 50 μm to 150 μm.

A value of a converter parameter θ may be less than or equal to 300pm²/V. θ is a size in the z-direction of the prepackage gap divided by amaximum electric field strength in the prepackage gap. Accordingly, theconverter parameter θ may be expressed as follows:

$\begin{matrix}{\theta = \frac{G_{1}}{E_{1}}} & (2)\end{matrix}$

In this equation, G₁ is the size of the prepackage gap in thez-direction, and E₁ is the maximum electric field strength in theprepackage gap. θ may be greater than or equal to 0.1 pm²/V. θ may beless than or equal to 250 pm²/V. θ may be in the range 2.0 pm²/V to 20pm²/V, in the range 3.0 pm²/V to 10 pm²/V, in the range 0.5 pm²/V to 100pm²/V, or in the range 2.0 pm²/V to 50 pm²/V.

The maximum electric field strength in the prepackage gap may be greaterthan or equal to 1 kV/mm, greater than or equal to 3 kV/mm, greater thanor equal to 5 kV/mm, or greater than or equal to 10 kV/mm. The maximumelectric field strength in the prepackage gap may be less than or equalto 50 kV/mm, less than or equal to 40 kV/mm, or less than or equal to 25kV/mm. The maximum electric field strength in the prepackage gap may bein the range 5 kV/mm to 40 kV/mm, in the range 10 kV/mm to 25 kV/mm, orin the range 3 kV/mm to 25 kV/mm.

The power electronics converter may further include a heat sink forremoving heat from power semiconductor prepackages. The heat sink may bespaced apart in the z-direction from the multi-layer planar carriersubstrate so as to define a gap (referred to herein as the heat sinkgap) between the multi-layer planar carrier substrate and the heat sink.The size in the z-direction of the heat sink gap between the multi-layerplanar carrier substrate and the heat sink may be less than or equal to10 mm, less than or equal to 5 mm, less than or equal to 3 mm, less thanor equal to 2.5 mm, less than or equal to 1 mm, or less than or equal to0.3 mm. The size of the heat sink gap may be greater than or equal to0.1 mm, greater than or equal to 0.5 mm or greater than or equal to 1mm. The size of the heat sink gap may be within the range 0.5 mm to 3mm, in the range 0.5 mm to 2.5 mm, in the range 0.5 mm to 2 mm, in therange 1 mm to 2 mm, or in the range 1.3 mm to 1.7 mm.

A value of a converter parameter φ may be less than or equal to 20nm²/V. φ is a size in the z-direction of the heat sink gap divided bythe maximum electric field strength in the heat sink gap. Accordingly,the converter parameter φ may be expressed as follows:

$\begin{matrix}{\varphi = \frac{G_{2}}{E_{2}}} & (3)\end{matrix}$

In this equation, G₂ is the size of the heat sink gap in thez-direction, and E₂ is the maximum electric field strength in the heatsink gap. φ may be less than or equal to 15 nm²/V. φ may be greater thanor equal to 0.01 nm²/V. φ may be in the range 0.25 nm²/V to 2.5 nm²/V,in the range 0.5 nm²/V to 1.5 nm²/V, in the range 0.02 nm²/V to 10nm²/V, or in the range 0.05 nm²/V to 5 nm²/V.

The maximum electric field strength in the heat sink gap may be greaterthan or equal to 0.1 kV/mm, greater than or equal to 0.2 kV/mm, orgreater than or equal to 1 kV/mm. The maximum electric field strength inthe heat sink gap may be less than or equal to 20 kV/mm, less than orequal to 15 kV/mm, less than or equal to 10 kV/mm, or less than or equalto 5 kV/mm. The maximum electric field strength in the heat sink gap maybe in the range 0.2 kV/mm to 10 kV/mm, in the range 1 kV/mm to 2 kV/mm,or in the range 1.3 kV/mm to 1.7 kV/mm.

Those skilled in the art will appreciate that the maximum electric fieldstrength in the prepackage gap and/or the heat sink gap may bedetermined using a mathematical and/or computational simulation method(e.g., Finite Element Analysis).

The maximum electric field strengths E₁, E₂ are maximum homogenous fieldstrengths in the respective gaps. In other words, the maximum fieldstrengths may exclude highly localized maxima such as singularities thatoccur at or near sharp edges. The maximum electric field strengths maybe determined at a point or in a region in the respective gap, where thepoint or region is located away in the x-y direction from a singularity(e.g., an edge or boundary) of the multi-layer planar carrier substrate,power semiconductor prepackage, and/or heat sink. The distance in thex-y direction between the point or region and the singularity (e.g., theedge of the multi-layer planar carrier substrate, the powersemiconductor prepackage, and/or the heat sink) may be defined by anumber of mesh cells of the simulation method (e.g., three mesh cells).By determining the maximum electric field strength in the point orregion, the maximum electric field strength is determined in a region ofthe electric field that is relatively homogenous.

The maximum electrical field strengths may be determined in a regionbetween two opposed and substantially parallel surface regions.

Where a prepackage gap is present, at least a portion of the prepackagegap may be filled with an electrically insulating material. Theelectrically insulating material, which may be a resin (e.g., adielectric resin or polymer resin) or another suitable insulatingmaterial, may have a plurality of voids within the region (e.g., volume)the electrically insulating material occupies.

A converter parameter σ may be greater than or equal to 10/mm. σ isdefined as an insulation fill factor of the electrically insulatingmaterial divided by a maximum void size of the plurality of voids.Accordingly, the σ may be expressed as follows:

$\begin{matrix}{\sigma = \frac{F}{R_{\max}}} & (4)\end{matrix}$

In this equation, F is the insulation fill factor and R_(max) is themaximum void size of the plurality of voids. The insulation fill factoris defined as a cumulated volume of the plurality of voids (e.g., ‘voidvolume’), subtracted from a volume of the electrically insulatingmaterial, divided by the volume of the electrically insulating material.This may be expressed as follows:

$\begin{matrix}{F = \frac{V_{IM} - V_{V}}{V_{IM}}} & (5)\end{matrix}$

In this equation, V_(IM) is the volume of the electrically insulatingmaterial and V_(V) is the cumulated volume of the plurality of voids.Hence, the converter parameter σ may also be expressed as:

$\begin{matrix}{\sigma = \frac{V_{IM} - V_{V}}{V_{V} \times R_{\max}}} & (6)\end{matrix}$

σ may be greater than or equal to 15/mm, greater than or equal to 18/mm,greater than or equal to 50/mm, greater than or equal to 80/mm, orgreater than or equal to 80/mm. σ may be less than or equal to 1000/mm,less than or equal to 800/mm, less than or equal to 500/mm, less than orequal to 200/mm, or less than or equal to 150/mm. σ may be in the range30/mm to 200/mm or in the range 50/mm to 150/mm.

The insulation fill factor may be greater than or equal to 90%, greaterthan or equal to 95%, greater than or equal to 99%, or greater than orequal to 99.99%.

The electrically insulating material may be described as an underfillmaterial of the semiconductor prepackage. In contrast to someapplications of underfill material in the field of electronics, whichprovide improved mechanical properties (e.g., stiffness), the underfillmaterial according to the present disclosure additionally oralternatively has an electrically insulating function to resist the highelectric fields developed in the power dense converters of the presentdisclosure.

The maximum void size of the plurality of voids may be less than orequal to 100 μm, less than or equal to 50 μm, less than or equal to 20μm, less than or equal to 10 μm, less than or equal to 5 μm, or lessthan or equal to 1 μm.

Those skilled in the art will appreciate that the term “voids” refers tothe inclusion of ‘foreign’ material, different to the electricallyinsulating material, in the region occupied by the electricallyinsulating material. The voids may be in solid (e.g., particle-like),liquid, or gaseous form. Example void materials may include soldermaterial, solder flux residues, air, washing liquid, and the like. Voidsmay be unintentionally introduced during the manufacturing process.

The maximum void size may be defined as the diameter of an equivalentspherical body having the same volume as the largest void. The largestvoid and/or the maximum void size may be determined by methods known tothose skilled in the art; for example, the largest void and/or themaximum void size may be determined by statistical methods determiningthe void size of a representative number of voids. The representativenumber of voids is smaller than the total number of the plurality ofvoids in the electrically insulating material.

A converter parameter τ may be less than or equal to 10,000 V. τ is aproduct of a dielectric strength of the electrically insulating materialand the maximum void size of the plurality of voids. Accordingly, τ maybe expressed as:τ=D×R _(max)  (7)

In this equation, D is the dielectric strength of the electricallyinsulating material and R_(max) is the maximum void size of theplurality of voids. τ may be less than or equal to 1,000 V, less than orequal to 500 V, or less than or equal to 250 V. τ may be greater than orequal to 1 V, greater than or equal to 10 V, greater than or equal to100 V, or greater than or equal to 150 V. τ may be in the range 100 V to300 V or in the range 150 V to 250 V.

The dielectric strength of the electrically insulating material, D, maybe greater than or equal to 1 kV/mm, greater than or equal to 10 kV/mm,or greater than or equal to 15 kV/mm. D may be less than or equal to 250kV/mm, less than or equal to 200 kV/mm, less than or equal to 100 kV/mm,or less than or equal to 50 kV/mm. D may be in the range 10 kV/mm to 30kV/mm or range 15 kV/mm to 25 kV/mm.

One or more (e.g., each) power semiconductor prepackage may furtherinclude one or more electrically conductive layers. The one or moreelectrically conductive layers may extend in the x-y direction. At leastone of the one or more electrically conductive layers may be embeddedwithin the solid electrically insulating material of the powersemiconductor prepackage. At least one of the one or more electricallyconductive layers may be located on an opposite side of the powersemiconductor switching element to the electrical connection side of theprepackage and connect to at least one of the terminals of the powersemiconductor switching element. At least one electrical connection mayextend in the z-direction from an electrically conductive layer of theprepackage to the electrical connection side of the prepackage.

Each power semiconductor prepackage may further include an electricalisolation layer. The electrical isolation layer may be located on anopposite side of the power semiconductor switching element to theelectrical connection side of the prepackage. The electrical isolationlayer may be of a ceramic material. The electrical isolation layer maybe embedded within the solid electrically insulating material of thepower semiconductor prepackage. The electrical isolation layer mayextend in the x-y direction.

The at least one capacitor may be connected to the at least one powerelectronics switching element through at least one of the conductivelayers of the multi-layer planar carrier substrate.

The at least one capacitor may be a ceramic capacitor.

The gate driver circuit may be electrically connected to the gateterminal of each power semiconductor switching element by one or moreelectrical connections extending in the z-direction.

The power electronics converter may further include a heat sink forremoving heat from power semiconductor prepackages.

The converter may further include a thermal interface layer (TIL)between a heat removal side of the prepackage and the heat sink. Theheat removal side of the prepackage is opposite the electricalconnection side of the prepackage.

The at least one prepackage may be located between the multi-layercarrier substrate and the heat sink. The heat sink may include one ormore recessed regions defining one or more chambers for receiving theprepackages. Adjacent chambers may be separated by a wall.

The at least one prepackage may be embedded within the multi-layercarrier substrate. The heat sink may be located adjacent a heat removalside of the prepackage opposite the electrical connection side of theprepackage.

A converter parameter n may be greater than or equal to 100 kW/m³K. η isa heat transfer coefficient between the heat removal side of the powersemiconductor prepackage and a cooling medium of the heat sink dividedby the size in the z-direction of the gap between the heat removal sideof the power semiconductor prepackage and the heat sink. Accordingly, ηmay be expressed as:

$\begin{matrix}{\eta = \frac{h}{G_{3}}} & (8)\end{matrix}$

In this equation, h is the heat transfer coefficient between the heatremoval side of the power semiconductor prepackage and the coolingmedium of the heat sink, and G₃ is the size in the z-direction of thegap between the heat removal side of the power semiconductor prepackageand the heat sink. The converter parameter η may be greater than orequal to 500 kW/m³K, greater than or equal to 1 MW/m³K, or greater thanor equal to 10 MW/m³K. The converter parameter η may be less than orequal to 1000 MW/m³K, less than or equal to 500 MW/m³K, less than orequal to 150 MW/m³K, or less than or equal to 100 MW/m³K. The converterparameter η may be in the range 1 MW/m³K to 1000 MW/m³K, in the range 10MW/m³K to 100 MW/m³K, in the range 20 MW/m³K to 50 MW/m³K, in the range125 kW/m³K to 75 MW/m³K, in the range 30 MW/m³K to 45 MW/m³K, or in therange 35 MW/m³K to 40 MW/m³K.

The heat transfer coefficient between the heat removal side of the powersemiconductor prepackage and a cooling medium of the heat sink, h, maybe greater than or equal to 0.1 kW/m²K, greater than or equal to 1kW/m²K, or greater than or equal to 5 kW/m²K. h may be less than orequal to 50 kW/m²K, less than or equal to 30 kW/m²K, or less than orequal to 20 kW/m²K. h may be in the range 2.5 kW/m²K to 15 kW/m²K or inthe range 5 kW/m²K to 10 kW/m²K. The size in the z-direction of the gapbetween the heat removal side and the heat sink, G₃, may be less than orequal to 2 mm, less than or equal to 1 mm, less than or equal to 0.8 mm,or less than or equal to 0.5 mm. G₃ may be greater than or equal to 0.05mm, greater than or equal to 0.1 mm, or greater than or equal to 0.2 mm.

The thermal interface layer (TIL) may have a thermal conductivity and amechanical compressibility.

A converter parameter Ω may satisfy 0.1 MNK/Wm<Ω<1 GNK/Wm. Ω is themechanical compressibility of the thermal interface layer divided by thethermal conductivity of the thermal interface layer. Accordingly, Ω maybe expressed as:

$\begin{matrix}{\Omega = \frac{M}{k}} & (9)\end{matrix}$

In this equation, M Is the mechanical compressibility of the thermalinterface layer and k is the thermal conductivity of the thermalinterface layer. Ω may be greater than or equal to 0.2 MNK/Wm, greaterthan or equal to 0.4 MNK/Wm, or greater than or equal to 0.6 MNK/Wm. Ωmay be less than or equal to 500 MNK/Wm, less than or equal to 100MNK/Wm, less than or equal to 10 MNK/Wm, or less than or equal to 5MNK/Wm. Ω may be in the range 0.25 MNK/Wm to 2 MNK/Wm, in the range 0.7MNK/Wm to 1.5 MNK/Wm, in the range 0.7 MNK/Wm to 1.5 MNK/Wm, or in therange 0.8 MNK/Wm to 0.9 MNK/Wm.

The mechanical compressibility of the thermal interface layer, M, may beless than or equal to 3000 MN/m² (300 MPa), may be less than or equal to100 MN/m² (10 MPa), or may be less than or equal to 10 MN/m² (1 MPa). Mmay be greater than or equal to 0.5 MN/m² (0.05 MPa). M may be in therange 1 MN/m² (0.1 MPa) to 5 MN/m² (0.5 MPa) or in the range 2.5 MN/m²(0.25 MPa) to 3.5 MN/m² (0.35 MPa).

The thermal conductivity of the thermal interface layer, k, may be lessthan or equal to 100 W/mK, less than or equal to 25 W/mK, less than orequal to 10 W/mK, or less than or equal to 5 W/mK. k may be greater thanor equal to 0.5 W/mK or greater than or equal to 1 W/mK. k may be in therange 2 W/mK to 10 W/mK or in the range 3 W/mK to 4 W/mK.

The thickness (e.g., size in the z-direction) of the thermal interfacelayer may be in the range 0.05 mm to 3 mm, in the range 0.075 mm to 1.5mm, in the range 0.1 mm to 0.75 mm, or in the range 0.15 mm to 0.25 mm.

An electrical isolation layer may be arranged between the one or morepower semiconductor switching elements of the prepackage and the heatsink. The electrical isolation may form part of the power prepackage orbe between the prepackage and the heat sink. The electrical isolationlayer may be arranged between the prepackage and the thermal interfacelayer. The electrical isolation layer may be arranged between thethermal interface layer and the heat sink.

A converter parameter ρ may be greater than or equal to 5 MVW/m²K. ρ isa product of a thermal conductivity of the thermal interface layer and abreakdown electrical field strength of the electrical isolation layer.Accordingly, ρ may be expressed as:ρ=k×E _(Break)  (10)

In this equation, k is the thermal conductivity of the thermal interfacelayer and E_(Break) is the breakdown electric field strength (which mayalso be referred to as a dielectric strength) of the electricalisolation layer. ρ may be greater than or equal to 10 MVW/m²K, greaterthan or equal to 25 MVW/m²K, or greater than or equal to 50 MVW/m²K. ρmay be less than or equal to 25 GVW/m²K, less than or equal to 5GVW/m²K, less than or equal to 500 MVW/m²K, or less than or equal to 250MVW/m²K. ρ may be in the range 25 MVW/m²K to 5 GVW/m²K or in the range50 MVW/m²K to 250 MVW/m²K.

The breakdown electric field strength (e.g., dielectric strength),E_(Break), of the electrical isolation layer may be greater than orequal to 5 kV/mm. E_(Break) may be less than or equal to 250 kV/mm.E_(Break) may be in the range 10 kV/mm to 50 kV/mm, in the range 10kV/mm to 100 kV/mm, or in the range 15 kV/mm to 25 kV/mm.

The thickness (e.g., size in the z-direction) of the thermal interfacelayer may be in the range 0.05 mm to 3 mm, in the range 0.075 mm to 1.5mm, in the range 0.1 mm to 0.75 mm, or in the range 0.15 mm to 0.25 mm.

The thickness (e.g., size in the z-direction) of the electricalisolation layer may be in the range 0.025 mm to 2 mm, in the range 0.025mm to 1 mm, in the range 0.05 mm to 1 mm, in the range 0.1 mm to 0.8 mm,or in the range 0.2 mm to 0.3 mm.

The thermal interface layer may have a relatively low electricalconductivity (e.g., a high dielectric strength) so as to reduce thedependence of the converter on additional electrical isolation measuressuch as the inclusion of the dedicated electrical isolation layerdescribed above. Hence, some embodiments may not include an electricalisolation layer. In such embodiments, a converter parameter λ may begreater than or equal to 1 TW/SK, λ being defined as a thermalconductivity of the thermal interface layer divided by an electricalconductivity of the thermal interface layer. The converter parameter λmay be expressed as:

$\begin{matrix}{\lambda = \frac{k}{P}} & (11)\end{matrix}$

In this equation, k is the thermal conductivity of the thermal interfacelayer and P is the electrical conductivity of the thermal interfacelayer. λ may be less than or equal to 100 PW/SK, less than or equal to10 PW/SK, less than or equal to 1 PW/SK, or less than or equal to 500TW/SK. λ may be greater than or equal to 10 TW/SK, greater than or equalto 50 TW/SK, greater than or equal to 100 TW/SK, or greater than orequal to 200 TW/SK. λ may be in the range 100 TW/SK to 500 TW/SK or inthe range 300 TW/SK to 400 TW/SK.

The electrical conductivity of the thermal interface layer, P, may beless than or equal to 0.1 pS/m (e.g., 1×10⁻¹³ S/m). P may be greaterthan or equal to 1 fS/m (e.g., 1×10⁻¹⁵ S/m). P may be in the range 5fS/m to 50 fS/m or in the range 7.5 fS/m to 25 fS/m.

The thermal conductivity of the thermal interface layer, k, may begreater than or equal to 0.1 W/mK. k may be less than or equal to 150W/mK. k may be in the range 0.5 W/mK to 100 W/mK, in the range 1 W/mK to25 W/mK, in the range 2 W/mK to 10 W/mK, or in the range 2.5 W/mK to 5W/mK.

The multi-layer planar carrier substrate may include at least one outerconductive layer and/or at least one internal conductive layer. The atleast one inner conductive layer may be thicker than (e.g., at leasttwice or at least three times as thick as) the outer conductive layer.The outer conductive layer and the inner conductive layer may beelectrically connected by at least one connection (e.g., a plurality ofconnections) extending in the z-direction through the carrier substrate.Each power semiconductor prepackage may further include an electricalconnection from at least one terminal of the one or more powersemiconductor switching elements to the outer conductive layer of themulti-layer planar carrier substrate. The electrical connection extendsin the z-direction through the solid insulating material.

Each power semiconductor prepackage may further include an electricalconnection from at least one of its terminals to an electricalconnection side of the power semiconductor prepackage. The electricalconnection extends in the z-direction through the solid insulatingmaterial. Electrical connections may extend from each of the terminalsto the electrical connection side of the power semiconductor prepackage,each electrical connection extending in the z-direction through thesolid insulating material.

At least one of the terminals of each of the at least one powersemiconductor switching elements may be connected to at least one of theconductive layers of the multi-layer planar carrier substrate at theelectrical connection side of the power semiconductor prepackage. The atleast one of the terminals may be a source and/or drain terminal of thepower semiconductor switching element.

For each power semiconductor prepackage, the electrical connection sideof the prepackage may form a flat surface. The prepackage may be surfacemounted at its electrical connection side to a planar surface of themulti-layer planar carrier substrate.

For each power semiconductor prepackage, each electrical connectionextending from the at least one of the terminals of the powersemiconductor switching element through the solid insulating materialmay terminate at the flat surface of the prepackage. Each powersemiconductor prepackage may be surface mounted to the surface of themulti-layer planar carrier substrate by soldering, sintering, or gluingof the terminated electrical connection to an electrical connection ofthe multi-layer planar carrier substrate.

The flat surface of the electrical connection side may further includeconductive pads (e.g., solder pads) for connecting the terminals of thepower semiconductor switching element to the surface of the multi-layerplanar carrier substrate.

The terminated electrical connection may be connected (e.g., soldered,sintered, or glued) to an external conductive layer of the multi-layercarrier substrate. These connections space apart each powersemiconductor prepackage from the planar surface of the multi-layerplanar carrier substrate to define a gap (referred to herein as theprepackage gap) between the multi-layer planar carrier substrate and theelectrical connection side.

The power electronics converter may be an AC-DC converter (e.g., aninverter or a rectifier). The AC-DC converter may be a multi-phase AC-DCconverter. For each phase of the plurality of phases, the power circuitmay include a phase leg. In this case, the parasitic inductance of thepower circuit of the commutation cell is the parasitic inductance of onephase leg. Those skilled in the art will understand that, aside frominherent variation due to variation in components and manufacture, theparasitic inductance of each phase leg is the same.

The AC-DC converter may be a two-level converter including two logicalswitches per phase. The number of power semiconductor prepackages perlogical switch may be greater than or equal to three. In some examples,a multi-phase (e.g., three-phase or four-phase) AC-DC converter includesgreater than 50 power semiconductor prepackages.

The power electronics converter may be a DC-DC converter. The powercircuit of the commutation cell may further include an inductor.

The at least one capacitor may be a single capacitor (e.g., a single“DC-link” capacitor or “input capacitor”). In other examples, a morecomplex DC filter including a plurality of capacitors may be used.

The electrical machine may be of any suitable type and configuration. Inone specific example, the electrical machine is a transversal flux(e.g., transverse flux) electrical machine. The transversal fluxelectrical machine may be air-cooled. The transversal flux electricalmachine and the power semiconductor prepackages may both be air-cooledby a common cooling system.

An electrical propulsion unit (EPU) for an aircraft is also provided.The EPU includes an electric motor and an AC-DC power electronicsconverter in accordance the AC-DC power electronics converters set outabove. The AC-DC power electronics converter is configured as aninverter and arranged to supply current to a winding of the electricmotor.

A gas turbine engine is also provided. The gas turbine engine includes aspool, an electrical machine having a rotor mechanically coupled to thespool, and an AC-DC power electronics converter in accordance the AC-DCpower electronics converters set out above. The AC-DC power electronicsconverter is arranged to supply current to or receive current from awinding of the electric machine.

An electrical power system for an aircraft is also provided. Theelectrical power system includes a power electronics converter inaccordance the power electronics converters set out above.

An aircraft is also provided. The aircraft includes the EPU, the gasturbine engine, or the electrical power system set out above. In onegroup of embodiments, the aircraft is an electric vertical take-off andlanding (eVTOL) aircraft having a plurality of EPUs as set out above.

The skilled person will appreciate that except where mutually exclusive,a feature described in relation to any one of the above aspects may beapplied mutatis mutandis to any other aspect. Further, except wheremutually exclusive, any feature described herein may be applied to anyaspect and/or combined with any other feature described herein.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will now be described by way of example only with referenceto the accompanying drawings, which are purely schematic and not toscale, and in which:

FIG. 1 is circuit diagram of one embodiment of a commutation cell of aone-phase, two-level AC-DC converter;

FIG. 2A is a schematic cross-section of the one-phase, two-level AC-DCconverter of FIG. 1 ;

FIG. 2B is a further schematic cross-section of the one-phase, two-levelAC-DC converter of FIG. 1 illustrating further properties such aselectric fields;

FIGS. 3A and 3B are schematic cross-sectional and plan views of theone-phase, two-level AC-DC converter of FIG. 2A, further illustratingthe calculation of the commutation cell volume;

FIG. 4A is a schematic cross-section of a single power semiconductorprepackage, showing details not visible in FIGS. 2A and 3A;

FIG. 4B is a schematic cross-section of the power semiconductorprepackage of FIG. 4A sandwiched between a multi-layer carrier substrateand a heat sink;

FIG. 4C is a schematic cross-section of an embodiment of a powersemiconductor prepackage sandwiched between a multi-layer carriersubstrate and a heat sink, and including an electrically insulatingmaterial in a prepackage gap;

FIG. 5A illustrates an embodiment of how low- and high-side powersemiconductor switching elements of a half-bridge may be electricallyconnected through a multi-layer carrier substrate;

FIG. 5B illustrates an embodiment of how power semiconductor switchingelements may be electrically connected in parallel through a multi-layercarrier substrate;

FIGS. 6A and 6B illustrate how the one-phase arrangements of FIGS. 2A,2B, 3A, 3B, 4A-C, 5A, and 5B may be extended to multiple phases;

FIG. 7A is a further schematic cross-sectional illustration of the powerelectronics converter of FIG. 2A;

FIG. 7B is a schematic cross-sectional illustration of an alternativeexemplary arrangement in which power semiconductor prepackages and aheat sink are arranged on opposite sides of the multi-layer carriersubstrate;

FIGS. 8A and 8B illustrate alternative embodiments of arrangements inwhich power semiconductor prepackages are embedded within a multi-layercarrier substrate;

FIG. 9A illustrates one embodiment of a two-level, three phase inverterpowering a three-phase motor;

FIG. 9B illustrates a plurality of H-bridge inverter circuits powering afour-phase motor, according to an embodiment;

FIG. 9C illustrates one embodiment of a two-level, three phase inverterwith a more complex DC-side filter;

FIG. 10A illustrates one embodiment of a DC-DC converter connected withthe terminals of a battery;

FIG. 10B illustrates one embodiment of a DC-AC-DC converter connectedwith the terminals of a battery;

FIG. 11A is a schematic illustration of one embodiment of an electricaircraft power and propulsion system;

FIG. 11B is a schematic illustration of one embodiment of amulti-channel electric power and propulsion system for an aircraftincluding six electrically powered propulsors;

FIG. 12A is a perspective view of one embodiment of an electric verticaltake-off and landing (eVTOL) aircraft including six electrically poweredpropulsors in an eVTOL flight configuration;

FIG. 12B is a perspective view of the eVTOL aircraft of FIG. 2A in aforward flight configuration;

FIG. 13 shows an arrangement of one embodiment of a turbofan engine foran aircraft;

FIG. 14A is a schematic illustration of one embodiment of a hybridelectric aircraft propulsion system; and

FIG. 14B illustrates one embodiment of an electrically powered propulsorsuch as may be used in an electric or hybrid electric propulsion system.

DETAILED DESCRIPTION

FIG. 1 is a circuit diagram of a power electronics converter 10. Forsimplicity and ease of explanation, one phase of a two-level AC-DCinverter is illustrated. The present disclosure is not, however, limitedto converters of this circuit type, and further examples are disclosedherein.

The power electronics converter 10 has a commutation cell having twoparts: a power circuit and a gate driver circuit.

The power circuit has two DC inputs (DC-IN_(L) and DC-IN_(H)) and asingle-phase AC output (AC-OUT). Connected between the DC inputs and theAC output of the power circuit is a half-bridge circuit including twopower semiconductor switching elements 121L, 121H. The letters ‘L’ and‘H’ designate the low- and high-voltage sides of the half-bridge whichare connected to the low- and high-voltage DC inputs. The power circuitalso includes an intermediate smoothing capacitor 14 (e.g., referred toas the ‘DC-link capacitor’ in the context of an AC-DC converter). Thefunction of the DC-link capacitor will be familiar to those skilled inthe art.

Each power semiconductor switching element 121L, 121H has threeterminals. In the case of a MOSFET, the terminals are referred to as thesource (S), the drain (D), and the gate (G). Current flowing from the DCinputs to the AC output passes between the source (S) and the drain (D),while the voltage and current at the gate (G) controls whether or notthe path between the source (S) and the drain (D) conducts. The powersemiconductor switching elements may be MOSFETs (e.g., Silicon Carbide(SiC) based MOSFETs), though other semiconductor technologies such asGalium Nitride (GaN) may be used. As will be understood by those skilledin the art, the use of MOSFETs may allow for the omission of discreteparallel-connected diodes due to the inherent ‘body diode’ character ofa MOSFET.

The gate driver circuit 13L, 13H is electrically connected to andconfigured to supply switching signals to the gate terminal of theMOSFET 121L, 121H to control the conduction of the MOSFETs (e.g., tocontrol whether current may flow between the source and drain terminalsor whether the flow of current is blocked). The gate driver circuiteffectively acts as an amplifier of signals received from a controller(not shown) (e.g., a digital controller that operates at and suppliessignals having lower voltages, such as 3 V to 5 V). In this example, thegate driver circuit is also connected to the drain terminals of theMOSFETs 121L, 121H, though those skilled in the art will appreciate thisis not necessarily the case and that the gate and drain terminals may beisolated from each other.

The power circuit is further shown to include an inductor L_(P). Theinductor L_(P) is not a discrete component of the power circuit, butinstead represents the combined parasitic inductance of the powercircuit. Parasitic inductance is the inherent inductance of componentsand the connections between components that is not intentionallyintroduced into the circuit. The gate driver circuits 13L, 13H are alsoshown to include inductors L_(G); these too represent the parasiticinductances of the gate driver circuits 13L, 13H and are not discretecomponents.

Parasitic inductance is a notable problem in power electronicsconverters because parasitic inductance creates a loss mechanism:switching losses. The higher the parasitic inductance, the higher theswitching losses. The magnitude of the switching losses also increaseswith the operating voltage of the power semiconductor switches 121L,121H and with the switching frequency of the power semiconductorswitches 121L, 121H. This provides that a high parasitic inductance alsolimits a system designer's ability to select higher values of theconverter operating voltage and maximum switching frequency, as theseare to be kept lower to keep the switching losses at a tolerable level.These are undesirable limitations. The use of a lower voltagenecessitates the use of a higher current to achieve the same power(P=I×V), which increases resistive losses (e.g., I²R losses) in thepower circuit and in, for example, electric machine windings to whichthe converter is connected. The use of a lower switching frequencylimits the quality of the output voltage/current waveform, which leadsto undesirable effects such a torque ripple in the rotor of anelectrical machine connected to the power electronics.

The present disclosure provides power electronics converters withcommutation cells having reduced parasitic inductances. As well asreducing the switching losses, this allows the adoption of increasedoperating voltages, increased switching frequencies, and higher voltageand current ramp rates during switching. Overall, this provides for asignificant increase in the operating efficiency of the converterscompared with state-of-the-art power electronics converters.

Table 2 provides exemplary values of the parasitic inductance of thepower circuit and the maximum operating efficiency of AC-DC convertersin the power range 50-400 kW. The powers quoted are peak rated powers(e.g., the highest electrical power that may be controlled by theconverter). This differs from the continuous rated power that dependson, for example, environmental conditions during operation and thecapabilities of the converter cooling system.

TABLE 2 Parasitic Inductance of Power Circuit, L_(P) (nH) Peak OperatingEfficiency State-of- Present State-of- Peak Rated the-Art Disclosurethe-Art Present Power (kW) Example Example Example Disclosure 50 40 895% to 96% 97% to >99% 100 20 4 95% to 96% 97% to >99% 150 15 3 95% to96% 97% to >99% 200 10 2 95% to 96% 97% to >99% 400 5 1 95% to 96% 97%to >99%

Only the parasitic inductance LP of the power circuit, and not of thewhole commutation cell, is quoted. This is because the gate drivercircuit is electrically decoupled from the power circuit, and so theparasitic inductances of the two circuits are not combined. It should,however, be appreciated that according to the present disclosure, theclose integration of the gate driver circuit within the commutation cellresults in a reduced value of L_(G).

As shown in Table 2, the value of parasitic inductance of powerelectronics converters according to the present disclosure may beapproximately five times lower than the comparable state-of-the-artexample. This, along with other measures disclosed herein, results inoperating efficiencies up to and in excess of 99%, which compares withvalues of 95-96% commonly achieved in state-of-the-art power electronicsmodules.

Table 2 also shows that the parasitic inductance of the power circuitmay decrease as the power rating increases. This is because the peakrated power may be increased through parallelization of the powersemiconductors of the power circuit (e.g., at higher power ratings eachlow-side MOSFET 121L of a given phase is implemented by multiple MOSFETsconnected in parallel). This parallel connection of the components hasthe additional effect of reducing the parasitic inductance of the powercircuit. While this provides a mechanism for reducing the parasiticinductance of the power circuit to any desired low value, the additionalcomponents significantly add to the weight and volume of the converter,reducing the power density.

Thus, a power electronics converter may be characterized by a converterinductance-volume parameter α, defined as the product of the parasiticinductance of the power circuit and the volume of the commutation cell:α=L _(P) ×Vol.  (12)

where:

L_(P)=Parasitic inductance of power circuit, and

Vol.=Smallest cuboidal volume enclosing commutation cell

As stated above, the volume is defined as the smallest cuboidal volumethat encloses the entire commutation cell (e.g., the combination of thepower circuit and the gate driver circuit). An example illustrating thecommutation cell volume is shown and will be described with reference toFIGS. 3A and 3B.

Table 3 shows exemplary values of a for power electronics convertersaccording to the present disclosure. The values of a arecharacteristically lower than the prior art and associated with acombination of high efficiency and high power density. Values of a arequoted in pHm³ (pico-Hm³, or ×10⁻¹² Hm³).

TABLE 3 α = L_(P) × Vol. Peak Rated (pHm³) Power (kW) Example 1 Example2 Example 3 50 0.60 1.70 4.80 100 0.42 1.20 3.40 150 0.37 1.04 2.94 2000.30 0.85 2.40 400 0.21 0.60 1.70

Example values of the parasitic inductance of the power circuit arequoted in Table 2 and may be less than or equal to 16 nH for convertershaving peak power ratings in the range 25-500 kW. The commutation cellvolume may increase with power rating, and values of less than 1,000 cm³may be provided for converters having powers up to 500 kW. Commutationcell volumes may be greater than or equal to 100 cm³, with volumes of150 cm³ to 600 cm³ striking a good balance between power density andrelative ease of heat removal.

For reference, Table 4 includes values of α_(p), which is thepower-normalized value of α, and values of the product of the power andparasitic inductance of the power circuit.

TABLE 4 α_(P) = α/P L_(P)*P Peak Rated (aHm³/W) (mHW) Power (kW) Example1 Example 2 Example 1 Example 2 50 12 96 0.2 0.8 100 4.2 34 0.2 0.8 1502.5 20 0.2 0.8 200 1.5 12 0.2 0.8 400 0.7 5.7 0.2 0.8

Returning to FIG. 1 , although a circuit diagram, the components of thecommutation cell are shown to be located on a carrier substrate 11. Thecarrier substrate 11 is a multi-layer carrier substrate 11 (e.g., arigid printed circuit board (PCB)) including alternating insulating andconductive layers that extend in the x-y plane. The components of thecommutation cell, including both the power circuit and gate drivercircuit, are mounted on the multi-layer planar carrier substrate 11 andare electrically connected through the multi-layer carrier substrate 11.The electrical connections are made through a combination of theconductive layers of the multi-layer carrier substrate 11 andconnections extending through the carrier substrate 11 in a z-directiondefined perpendicular to the x-y direction of the planar substrate andits conductive layers. The connections extending the z-direction may,for example, be conductive vias or filled holes (e.g., laser micro-viashaving diameters of the order of 100 μm), and may have an x- and/ory-component as well as z-component (e.g., the connections may beparallel to or form an angle with respect to the z-direction).

FIG. 1 also schematically illustrates that the power semiconductorswitching elements 121L, 121H of the power circuit are each included ina power semiconductor prepackage 12L, 12H. Those skilled in the art willunderstand this to provide that the power semiconductors (e.g., MOSFETs)are embedded in a solid insulating material that electrically isolatesthe power semiconductors and their terminals from their surroundings.The prepackages and their connections to each other and to othercomponents of the commutation cell will be described in more detailbelow.

FIG. 2A is a schematic cross-section of a power electronics converter 10in accordance with the present disclosure. Again, for clarity and easeof explanation, one phase of a two-level AC-DC converter is illustrated.The x-direction and z-direction are indicated. The y-direction extendsinto the plane of the page.

The converter 10 includes a multi-layer carrier substrate 11, powersemiconductor prepackages 12, a gate driver circuit 13, an intermediate(DC-link) capacitor 14, DC inputs (DC-IN), and an AC output (AC-OUT).FIG. 2A further illustrates an integrated heat sink 15 that interfaceswith a cooling side (also referred to herein as the heat removal side)of the prepackages 12 via a thermal interface layer 16.

The multi-layer carrier substrate 11 has opposed first and second planarsurfaces 111 a and 111 b that define an x-y direction and a z-directionperpendicular to the x-y direction. The multi-layer substrate 11includes alternating layers of insulating and electrically conductivematerial. The electrically conductive layers 112 may be formed ofcopper, though the electrically conductive layers 112 may be formed ofany suitable conductive material such as silver, gold, or aluminum. Theinsulating layers may be formed of the base material of the carriersubstrate 11.

The multi-layer carrier substrate 11 may be a rigid PCB, in which casethe base material and conductive layers may be a glass woven fabricimpregnated with resin, as is known in the art of PCB manufacture. Themulti-layer carrier substrate 11 may, however, take a different form(e.g., a ceramic-based carrier substrate or a flexible PCB havingflexible polymer film base). A rigid material may be used, partly sothat the carrier substrate 11 may effectively act as a structuralcomponent of the converter 10.

The number of layers in the multi-layer carrier substrate 11 may varybetween applications, partly depending on the specifics of the powercircuit (e.g., the number of phases in an AC-DC converter and the numberof power semiconductors connected in parallel in each logical switch).In one specific example, there are sixteen layers, including eightinsulating layers and eight conductive layers 112.

Referring to FIG. 4B, in some examples, at least a portion of one of theplanar surfaces 111 b of the carrier substrate 11 carries a metal layer1121 that defines one or more electrical connection regions on thesurface 111 b of the substrate 11. Referring again to FIG. 2A,additionally or alternatively to the outer layer, electrical connectionsin the form of, for example, vias or filled holes vias 114 extend in thez-direction through the carrier substrate 11 and terminate at the planarsurface 111 b. The points at which the electrical connections 114terminate define electrical connection points at the planar surface 111b of the carrier substrate 11.

Each prepackage 12 includes a power semiconductor switching element 121embedded in a solid insulating material 122. Embedding the powersemiconductor switching elements 121 in solid insulating materialprovides there are no air gaps surrounding the semiconductor chips andthe terminals, reducing the risk of electrical breakdown even where highconverter voltages are used. This allows higher voltages to be usedand/or for the power semiconductors 121 and other components to bespaced closer together, increasing the power density of the converter10.

Table 5 provides exemplary values of the maximum blocking voltage (e.g.,the source-drain blocking voltage or ‘rated voltage’) of the powersemiconductor switching elements 121 in accordance with the presentdisclosure.

TABLE 5 Power Semiconductor Blocking Voltage Peak Rated (V) Power (kW)Example 1 Example 2 Example 3 50 1200 800 1600 100 1200 800 1600 1501200 800 1600 200 1200 800 1600 400 1200 800 1600

As shown, the source-drain blocking voltages utilized in accordance withthe present disclosure are high. The blocking voltage may be in therange of 600 V to 1,800 V, and values greater than or equal to 800 V maybe provided to limit the peak current and reduce conduction losses.State of the art power electronics converters may have much lowerblocking voltages, with voltages of even 600 V being rare. As alsoshown, the blocking voltage does not increase with the peak power of theconverter. This is because most or all the increase in peak rated poweris achieved through parallelization in the power circuit. In otherexamples, a somewhat higher blocking voltage may be used for converterswith higher power ratings (e.g., higher than 200 kW).

Each power semiconductor switching element 121 may have at least threeterminals, including a gate terminal (G) for switching the conductionstate of the switching element 121. In some embodiments, the powersemiconductor switching element 121 may have more than three terminals(e.g., if one or more terminals are provided for measurement, such as aKelvin terminal) or if additional shorted terminals are present. Inembodiments, the power semiconductor switching elements 121 are MOSFETs,in which case the terminals are designated the source (S), gate (G), anddrain (D). In principle, however, other materials semiconductorswitching devices (e.g., IGBTs) may be used in place of MOSFETs. Thesolid insulating material 122 may be any suitable insulating material(e.g., FR4).

As will be described in more detail below with reference to FIGS. 4A-4B,electrical connections extend from the terminals of the powersemiconductor switching elements 121 through the solid insulatingmaterial 122 and terminate at an electrical connection side/surface 123a of the prepackage 12. Thus, the electrical connection side 123 a ofthe prepackage 12 forms a substantially flat surface with exposedelectrical connection points.

The electrical connection side of the prepackages 123 a faces one of theplanar surfaces 111 b of the carrier substrate 11. The prepackages 12are surface mounted at their flat electrical connection sides 123 a tothe planar surface 111 b, for example, by soldering, sintering, orgluing (e.g., sinter gluing using a mix of glue and sinter paste) theelectrical connection points of the prepackages 12 to the electricalconnection points or region(s) of the planar surface of the carriersubstrate 11. FIG. 2A schematically illustrates soldered/sintered/gluedconnections 113.

The thickness (e.g., the size in the z-direction) of the connections113, which define a prepackage gap between the opposed surfaces 123 a,111 b of the prepackages 12 and the carrier substrate 11, is small. Forexample, the thickness and gap may be, when measured parallel to thez-direction, less than 500 μm (e.g., between 20 μm and 250 μm). In aspecific embodiment, the gap is 100 μm.

Terminating the electrical connections from the chip terminals at a flatsurface 123 a of the prepackage 12 and using surface mounting to formthe onward electrical connections through the PCB reduces the overallsize of the converter 10 in the z-direction, which reduces the size andweight of the converter 10. Further, the surface mounting of theprepackages 12 reduces the impact of ‘open loop’ effects in theelectrical connections between the power semiconductors 121, the gatedriver circuit 13, and the DC-link capacitor 14. This may substantiallyreduce the parasitic inductance of the commutation cell, which reducesswitching losses and allows the use of, for example, an increasedswitching frequency.

The gate driver circuit 13 is electrically connected to and configuredto supply switching signals to the gate terminals of the powersemiconductor switching elements 121. In the illustrated embodiment, thegate driver circuit 13 is mounted to the first planar surface 111 a ofthe carrier substrate 11, opposite to the second planar surface 111 bthat faces the power semiconductor prepackages 12. In other embodiments,the prepackages 12 and the gate driver circuit 13 may be mounted at thesame side of the carrier substrate 11 (e.g., the second side 111 b). Inthe illustrated embodiment, the electrical connection 114 between thegate driver circuit 13 and the gate terminals of the power semiconductorswitching elements 121 extends in the z-direction through the carriersubstrate to the surface 111 b of the carrier substrate 11. The onwardpath then passes through a solder connection 113 and then through theelectrical connection that passes in the z-direction through the solidinsulating material 122 of the prepackages 12 to the gate terminal. Inother embodiments, the connection between the gate driver circuit 13 andthe gate terminals may be made through one or more conductive layers 112of the substrate 11.

Table 6 provides exemplary values of the switching frequency of thepower semiconductor switching elements of a power electronics converterin accordance with the present disclosure. Table 6 also includes maximumabsolute values of the rate of change of the source-drain voltage(measured in units of kilo-Volts per micro-second) of the powersemiconductor switching elements 121 during a switching cycle. Thequoted values are maximum values of the switching frequency to occurduring operation.

TABLE 6 f_(max) |dV/dt|_(max) Peak Rated (kHz) (kV/μs) Power (kW)Example 1 Example 2 Example 1 Example 2 50 50 80 30 45 100 50 80 30 45150 50 80 30 45 200 50 80 30 45 400 50 80 30 45

In many converter arrangements, the switching frequency of each powersemiconductor switching element 121 is the same and the same voltagevalue is used, such that the maximum values quoted above are the samefor each individual power semiconductor switching element 121. However,some converter architectures (e.g., multi-level converter architecturessuch as modular multi-level converter architectures) that interface withmultiple network voltage levels use different voltages and/or switchingfrequencies for different power semiconductor switching elements 121. Inthese cases, the quoted values correspond to the maximum value of anyindividual power semiconductor switching element 121 in the converter.

Converters according to the present disclosure use maximum frequenciesgreater than 10 kHZ, though switching frequencies in excess of 30 kHzmay be provided, which as discussed below, may facilitate a reduction inthe required capacitance in the power circuit. Unlike manystate-of-the-art systems in which the parasitic inductance of theconverter commutation cell limits the maximum usable frequency, the lowparasitic inductance of the converter commutation cell may provide othersystem constraints that limit the maximum frequency. For example, amaximum desirable switching frequency may be imposed by the capabilitiesof the insulation of the windings of an electrical machine thatinterfaces with an AC-DC converter. A switching frequency of less thanor equal to 100 kHz may be used.

The use of a high switching frequency and high source-drain blockvoltage results in a notably high value of the maximum the rate ofchange of the source-drain voltage during operation. Rapid switchingbetween the on- and off-states of the power semiconductor switchingelements results in clean switching and improved output waveforms, whichlimits the harmonic content of the waveforms and improves, for example,converter efficiency, and reduces electric machine torque ripple. In theabove examples, the maximum rate of change of the source-drain voltageduring operation is in the range 30 to 45 kV/μs. However, lower values(e.g., 10 to 20 kV/μs) or higher values (e.g., greater than 50 kV/μs oras high or higher than 100 kV/μs) may be used, for example, where arelatively low or high switching frequency is utilized.

Thus, a power electronics converter may be characterized by a converterswitching parameter β, defined as the product the maximum switchingfrequency of the switching signals and the maximum rate of change of thesource-drain voltage of the power semiconductor switching elementsduring operation.β=f _(max) ×|dv/dt| _(max)  (13)

where:

-   -   f_(max)=Maximum frequency of switching signals, and    -   |dV/dt|_(max)=Maximum rate of change of source-drain voltage        during operation

Table 7 shows exemplary values of β for power electronics convertersaccording to the present disclosure. The values of β arecharacteristically higher than the prior art and associated with acombination of high efficiency, high power density, and high-quality(e.g., highly sinusoidal AC) output waveforms. Values of β are quoted inPV/s² (Peta-V/s² or ×10¹⁵ V/s²).

TABLE 7 β = f_(max) × |dv/dt|_(max) Peak Rated (PV/s²) Power (kW)Example 1 Example 2 Example 3 50 0.5 1.5 5.0 100 0.5 1.5 5.0 150 0.5 1.55.0 200 0.5 1.5 5.0 400 0.5 1.5 5.0

Converters in accordance with the present disclosure may have a value ofβ that is greater than or equal to 0.3 PV/s². Values of β may be lessthan 10 PV/s² to mitigate against problems such as, for example,insulation breakdown. Values in the range 0.8 PV/s²≤β≤5 PV/s² or 1.0PV/s²≤β≤2.5 PV/s² may strike a good balance between the competingeffects.

Returning to FIG. 2A, the intermediate capacitor 14, which forms part ofthe power circuit of the commutation cell, is electrically connected tothe power semiconductor switching elements 121 through one or moreconductive layers 112 of the multi-layer carrier substrate 11. In theillustrated embodiment, the capacitor 14 is mounted to the first planarsurface 111 a of the carrier substrate 11, opposite to the second planarsurface 111 b that faces the power semiconductor prepackages 12. Inother embodiments, the prepackages 12 and the capacitor 14 may bemounted at the same side of the carrier substrate 11 (e.g., the secondside 111 b). Electrical connections between the capacitor 14 and theconductive layers 112 of the multi-layer carrier substrate 11 may beformed in a number of ways. In one example, the capacitor 14 is surfacemounted to the substrate 11 in a similar manner as the prepackages 12(e.g., by soldering, sintering, or gluing electrical contacts of thecapacitor 14 to an electrical connection region of the surface 111 a ofthe substrate 11). The electrical connection region may be an outerconductive layer on the surface 111 a of the substrate (similar to thelayer 1121 shown in FIG. 4B) or an exposed end of one or more vias orfilled holes 114 that extends in the z-direction from the surface 111 aof the substrate 11 to an internal conductive layer 112 of the substrate11. The connection between the capacitor 12 and the multi-layersubstrate 11 may be made through through hole technology (THT) as analternative to surface mount technology (SMT), both of which will befamiliar to those skilled in the art.

In state-of-the-art converters, the capacitor(s), which are an essentialcomponent of most AC-DC and DC-DC converter architectures, are asignificant contributor to both size and weight. This is a particularproblem in the context of aerospace applications, which are bothsensitive to size and weight and require relatively high powers (e.g.,compared to electric vehicles and domestic appliances), whichnecessitate a higher total rated power circuit capacitance. According tothe present disclosure, however, the reduction in parasitic inductanceallows for a lower capacitance per unit rated power. This is partlybecause the low parasitic inductance allows for a high switchingfrequency. Increasing the switching frequency may decrease the requiredcapacitance of the power circuit.

Table 8 provides exemplary values of the total rated capacitance of thepower circuit. Values of the total rated capacitance normalized by peakrated power of the converter are also included. It will be understoodfrom the above that the values of the total rated capacitance and thenormalized capacitance are low compared with state-of-the-art powerelectronics converters.

TABLE 8 Total Rated Capacitance C C/P Peak Rated (μF) (nF/W) Power (kW)Example 1 Example 2 Example 1 Example 2 50 15 40 0.3 0.8 100 30 80 0.30.8 150 45 120 0.3 0.8 200 60 160 0.3 0.8 400 120 320 0.3 0.8

Values of the total rated capacitance normalized by the peak rated powermay be less than 5 nF/W or less than 1 nF/W to achieve a low size andweight for the converter. The use of low values of the capacitance mayalso allow for the use of low-weight capacitor technologies (e.g.,ceramic capacitors), allowing for a further weight reduction.Capacitance is known to somewhat vary with operating conditions, andtherefore, the literature may quote nominal values of capacitance. InTable 8, the quoted capacitances are those measured at nominalconditions of 25° C. (298K) and 1,000 V DC, which is typical forcapacitance measurements.

The term “total rated capacitance of the power circuit” is the totalcapacitance of all the capacitors in the power circuit. In the simplestcases, there may be a single capacitor. For example, the one-phasetwo-level AC-DC converter circuit of FIG. 1 includes a single DC linkcapacitor, as does the three-phase, two-level AC-DC converter circuit ofFIG. 9A. In other examples, there may be multiple capacitors: there maybe a separate DC-link capacitor for each phase, as is the case in theH-bridge based arrangement of FIG. 9B and the three-phase, two-levelcircuit of FIG. 9C. Where there is parallelization of the powersemiconductors in the power circuit, each power semiconductor switchingelement may be associated with its own capacitor, or multipleparallel-connected power semiconductor switching elements 121 may beconnected with a single larger capacitor.

A power electronics converter may be characterized by a converterfrequency-capacitance parameter γ, defined as the total ratedcapacitance of the power circuit divided by the product of the peakrated power output of the power electronics converter and the maximumswitching frequency of the gate switching signals:

$\begin{matrix}{\gamma = \frac{C}{P \times f_{\max}}} & (14)\end{matrix}$

-   -   where:    -   C=Total rated capacitance of power circuit of commutation cell,    -   P=Peak rated power of power electronics converter, and    -   f_(max)=Maximum frequency of switching signals.

Table 9 shows exemplary values of γ for power electronics convertersaccording to the present disclosure. The values of γ arecharacteristically lower than the prior art and associated with acombination of high efficiency and high power density. Values of γ arequoted in fFs/W (femto-Fs/W or ×10⁻¹⁵ Fs/W).

TABLE 9 γ = C/(P × f_(max)) Peak Rated (fFs/W) Power, P (kW) Example 1Example 2 Example 3 50 2.5 10 100 100 2.5 10 100 150 2.5 10 100 200 2.510 100 400 2.5 10 100

Power electronics converters in accordance with the present disclosuremay have values of γ less than or equal to 150 fFs/W. γ may be greaterthan 1.0 fFs/W, with the provision of a lower bound limiting problemsassociated with, for example, insulation breakdown at high switchingfrequencies. Values of γ may be in the range 4.0 fFs/W≤γ≤25 fFs/W, whichmay strike a good balance between high power density and reliableoperation.

A power electronics converter may also be characterized by a converterfrequency-capacitance parameter δ, defined as the maximum source-drainblocking voltage of the power semiconductor switching elements of thepower circuit divided by the product of the parasitic inductance of thepower circuit and the total rated capacitance of the power circuit:

$\begin{matrix}{\delta = \frac{V_{block}}{L \times C}} & (15)\end{matrix}$

-   -   where:    -   V_(block)=Maximum source-drain blocking voltage of power        semiconductors,    -   L=Parasitic inductance of power circuit, and    -   C=Total rated capacitance of power circuit of commutation cell.

Table 10 shows exemplary values of δ for power electronics convertersaccording to the present disclosure. The values of δ arecharacteristically higher than the prior art and associated with acombination of high efficiency and high power density. Values of δ arequoted in PV/FH (Peta-V/FH or ×10⁻⁵ V/FH).

TABLE 10 Peak Rated $\delta = \frac{V_{block}}{L \times C}$ Power(PV/FH) (kW) Example 1 Example 2 Example 3  50 1.0 6.0 30 100 1.0 6.0 30150 1.0 6.0 30 200 1.0 6.0 30 400 1.0 6.0 30

Power electronics converters in accordance with the present disclosuremay have values of δ greater than 0.5 PV/FH. The value of δ may be lessthan or equal to 40 PV/s². The value of δ may be greater than or equalto 1.5 PV/s². The value of γ may be in the range 2.5 PV/s² to 25 PV/s².The value of γ may be in the range 4.0 PV/s² to 15 PV/s².

A power electronics converter may also be characterized by a converterfrequency-capacitance parameter c, defined as:

$\begin{matrix}{\varepsilon = \frac{f_{\max} \times {❘\frac{dv}{dt}❘}\max}{L \times C}} & (16)\end{matrix}$

where:

-   -   f_(max)=Maximum frequency of switching signals,    -   |dV/dt|_(max)=Maximum rate of change of source-drain voltage        during operation,    -   L=Parasitic inductance of power circuit, and    -   C=Total rated capacitance of power circuit of commutation cell.

Table 11 shows exemplary values of ε for power electronics convertersaccording to the present disclosure. The values of ε arecharacteristically lower than the prior art and associated with acombination of high efficiency, high power density, and good-qualityoutput waveforms. Values of c are quoted in units of ×10²⁷ V/s⁴.

TABLE 11 Peak Rated$\varepsilon = \frac{f_{\max} \times {❘\frac{dv}{dt}❘}\max}{L \times C}$Power (×10²⁷ V/s⁴) (kW) Example 1 Example 2 Example 3  50 1.9 7.5 30 1001.9 7.5 30 150 1.9 7.5 30 200 1.9 7.5 30 400 1.9 7.5 30

Power electronics converters in accordance with the present disclosuremay have values of ε less than or equal 10²⁹ V/s⁴. The value of ε may begreater than 5×10²⁶ V/s⁴, as lower values may be associated with, forexample, insulation failure, though this will depend to some extent onthe application requirements (e.g., if high quality insulation may beprovided in an electrical machine). Values of ε may be in the range1.5×10²⁷ V/s⁴ to 3×10²⁸ PV/s², as this strikes a good balance betweenpower density, efficiency, and reliability.

Returning to FIG. 2A, components of the converter 10, particularly thepower semiconductor switching elements 121, generate heat that is to beremoved. Heat removal may be a particularly notable consideration inpower electronics converters in accordance with the present disclosurebecause of their compactness and high power density. To this end, theconverter 10 includes an integral heat sink 15 that is in close thermalcontact with power semiconductor prepackages 12. To provide efficientheat removal, a solid path is provided between a heat removal side 123 bof the prepackages (e.g., the underside of the prepackage 12 asillustrated in FIG. 2 , opposite the electrical connection side 123 a)so that the heat sink 15 removes heat from the prepackages 12 by thermalconduction. In the illustrated example, a plurality of the prepackages12 of the converter 10, and optionally all of the prepackages 12 of theconverter 10, share a common heat sink 15.

The heat sink 15 itself may be of any suitable design. The heat sink 15may, for example, be formed of aluminum or another thermally conductivematerial, cooled by a cooling flow of a coolant that may be a gas (e.g.,air) or a liquid (e.g., water or an oil). In some embodiments, a surfaceof the heat sink 15 opposite the prepackages 12 may be subject to animpinging flow of coolant to increase the coefficient of heat transferbetween the prepackages 12 and the cooling medium of the heat sink 15.

To provide efficient heat removal by conduction, there is to be a goodquality and consistent thermal interface between the heat removal side123 b of each of the prepackages 12 and the heat sink 15. This may bechallenging because, although the heat removal side 123 b of theprepackages 12 may be configured to be flat, there does exist somemanufacturing and assembly tolerance. For example, the thickness of theelectrical contacts 113 may vary slightly between and within prepackages12, which may result in prepackage tilting and/or inconsistent distancesbetween the heat removal side 123 b and the heat sink 15. As anotherexample, the multi-layer carrier substrate 11 may bend or locallydeform. In view of this, the converter 10 is also shown to include athermal interface layer (TIL) 16 between the heat removal side 123 b ofthe prepackages 12 and the heat sink 15. The TIL 16 is included toprovide a good quality thermal interface between the prepackages 12 andthe heat sink 15. The TIL accommodates tolerance issues while alsoproviding a thermally conductive path in the z-direction between theheat removal side 123 b of each of the prepackages 12 and the heat sink15. In some examples, the TIL 16 may also have a high thermalconductivity on the x-y plane to spread heat across the surface of theheat sink 15. This may be of particular use where a single TIL 16 servesmultiple prepackages 12.

The TIL 16 may take one of a number of different forms, including solids(e.g., a solder layer, a foil, or a film), semi-solids (e.g., a paste),or a liquid. In one group of examples, the TIL 16 is a layer of solder(e.g., indium-tin solder). In this case, to provide a good qualitysolder connection, each prepackage 12 may have its own TIL 16 ratherthan a single TIL. In another group of examples, the TIL is a foil(e.g., indium-tin or graphene foil) that is both thermally conductiveand flexible. Utilizing a TIL with some compressibility may beadvantageous for both accommodating manufacturing tolerances andpreventing separation of the heat sink 15 from the prepackages 12during, for example, vibration. The TIL 16 may have a thermalconductivity of at least 1 W/mK or at least 2.5 W/mK. The thickness ofTIL 16 may be less than a few mm (e.g., less than 1 mm), and in onegroup of examples, the TIL has a thickness of between 100 μm and 500 μm.

A power electronics converter may be characterized by a converter heattransfer parameter η, defined as:

$\begin{matrix}{\eta = \frac{h}{G_{3}}} & (17)\end{matrix}$

-   -   where:    -   h=heat transfer coefficient from prepackage to cooling medium,        and    -   G₃=distance in z-direction between heat removal side of        prepackage and heat sink.

The gap G₃ is labelled in FIG. 2A. Table 12 shows exemplary values of ηfor power electronics converters according to the present disclosure.The values of η are characteristically high and associated with acombination of a compactness, high efficiency, and high power density.Values of η are quoted in units of MW/m³K.

TABLE 12 Peak Rated $\eta = \frac{h}{G_{3}}$ Power (MW/m³K) (kW) Example1 Example 2 Example 3  50 0.125 37.5 150 100 0.125 37.5 150 150 0.12537.5 150 200 0.125 37.5 150 400 0.125 37.5 150

Power electronics converters in accordance with the present disclosuremay have values of η greater than or equal 100 kW/m³K (0.1 MW/m³K).However, values of η greater than or equal 10 MW/m³K may be provided.

Table 13 shows exemplary values for the heat transfer coefficient h andthe size of the gap, G₃, between the heat removal side of the powersemiconductor prepackage and the heat sink.

TABLE 13 Peak h G₃ Rated (kW/m²K) (mm) Power Example Example ExampleExample Example Example (kW) 1 2 3 1 2 3 50 0.1 7.5 30 0.8 0.2 0.05 1000.1 7.5 30 0.8 0.2 0.05 150 0.1 7.5 30 0.8 0.2 0.05 200 0.1 7.5 30 0.80.2 0.05 400 0.1 7.5 30 0.8 0.2 0.05

A power electronics converter may also be characterized by a thermalinterface parameter, Ω, defined as:

$\begin{matrix}{\Omega = \frac{M}{K}} & (18)\end{matrix}$

where:

-   -   M=mechanical compressibility of thermal interface layer, and    -   k=thermal conductivity of thermal interface layer.

Table 14 shows exemplary values for the thermal interface parameter Ω.Values of Ω are quoted in units of MNK/Wm (Mega-NK/Wm, equal to 10⁶NK/Wm).

TABLE 14 Peak Rated $\Omega = \frac{M}{k}$ Power (MNK/Wm) (kW) Example 1Example 2 Example 3  50 0 0.86 10³ 100 0 0.86 10³ 150 0 0.86 10³ 200 00.86 10³ 400 0 0.86 10³

Converters of the present disclosure may have values of Ω that satisfy0.1 MNK/Wm<Ω2<1 GNK/Wm or 0.25 MNK/Wm<Ω<2 MNK/Wm. Thermal interfaceparameters in this range may provide a good combination of heat transferand mechanical properties.

Table 15 shows exemplary values for the mechanical compressibility, M,of the thermal interface layer as well as for the thermal conductivity kof the thermal interface layer.

TABLE 15 Peak M k Rated (MN/m²) (W/mK) Power Example Example ExampleExample Example Example (kW) 1 2 3 1 2 3 50 0 3 100 1 3.5 90 100 0 3 1001 3.5 90 150 0 3 100 1 3.5 90 200 0 3 100 1 3.5 90 400 0 3 100 1 3.5 90

FIG. 2B is a further illustration of the converter 10 of FIG. 2A andillustrates the electric fields developed between components of theconverter 10.

The prepackage gap, labelled G1 in FIG. 2B, is visible between thesecond surface 111 b of the multi-layer carrier substrate 11 and theelectrical connection side/surface 123 a of the prepackages 12. Theprepackage gap G1 has a size that may be measured in the z-direction. Afirst electric field 50 is established in the prepackage gap G1resulting from the potential difference (e.g., voltage) between thesecond surface 111 b of the multi-layer carrier substrate 11 and theelectrical connection surface 123 a of the prepackages 12.

A power electronics converter may be characterized by a converterparameter θ, defined as a size in the z-direction of the prepackage gapdivided by a maximum electric field strength in the prepackage gap(referred to herein as the first maximum electric field strength):

$\begin{matrix}{\theta = \frac{G_{1}}{E_{1}}} & (19)\end{matrix}$

where:

-   -   G₁=Size of prepackage gap in z-direction, and    -   E₁=Maximum electric field strength in prepackage gap.

Table 16 shows exemplary values for the converter parameter θ for powerelectronics converters according to the present disclosure. The valuesof θ are characteristically lower than the prior art and associated witha high power density. Values of θ are quoted in units of pm²/V(pico-m²/V, or ×10⁻¹² m²/V).

TABLE 16 Peak Rated Power$\theta = {\frac{G_{1}}{E_{1}}\left( {{pm}^{2}/V} \right)}$ (kW) Example1 Example 2 Example 3  50 0.5 6.25 250 100 0.5 6.25 250 150 0.5 6.25 250200 0.5 6.25 250 400 0.5 6.25 250

Power electronics converters in accordance with the present disclosuremay have values of θ less than or equal to 300 pm²/V. The value of θ maybe greater or equal to 0.1 pm²/V, as lower values may be associatedwith, for example, greater risk of electrical breakdown. Values of θ maybe in the range 2.0 pm²/V to 50 pm²/V, as this may strike a good balancebetween power density and reliability.

Those skilled in the art will appreciate that the first maximum electricfield strength is a maximum homogenous electric field strength. In otherwords, the first maximum electric field strength is the maximum fieldstrength determined in a location sufficiently spaced away from sharpedges and/or obstructions in the gap that may result in electric fieldsingularities or other highly localized maxima. For example, FIG. 2Bshows a first region 52 in the prepackage gap in which the first maximumelectric field strength may be determined, as the first region 52 isspaced from any edge or boundary of the multi-layer planar carriersubstrate or of the power semiconductor prepackage in the x-y direction.FIG. 2B shows that the first region 52 is spaced in the x-direction by afirst distance 54 from a prepackage edge 123 e. In a similar manner, thefirst region 52 is spaced apart from another prepackage edge 123 e inthe y-direction (not visible here, as the y-direction is perpendicularto the drawing plane). Such offset of the first region 52 in the x-ydirection towards the inside of the prepackage gap provides that themaximum field strength is determined over a suitably homogenous regionof the electric field. Where the maximum electric field strength isdetermined by modelling (e.g., finite element analysis (FEA)), themaximum electric field strength may be determined at a location at leastthree mesh cells from any singularity in the model (e.g., spikes, sharpedges, triple points).

Table 17 provides exemplary values for the size G₁ of the prepackage gapin the z-direction, as well as the maximum electric field strength E₁ inthe prepackage gap.

TABLE 17 Peak Rated G₁ (μm) E₁ (kV/mm) Power Example Example ExampleExample Example Example (kW) 1 2 3 1 2 3 50 20 100 250 1 16 40 100 20100 250 1 16 40 150 20 100 250 1 16 40 200 20 100 250 1 16 40 400 20 100250 1 16 40

The second surface 111 b of the multi-layer carrier substrate 11 and aheat sink 15 are spaced apart in the z direction in a substantiallyparallel manner so as to form a heat sink gap, which is labelled G2 inFIG. 2B. A second electric field 56 is established in the heat sink gapG2, resulting from the potential difference (e.g., voltage) between thesecond surface 111 b of the multi-layer carrier substrate 11 and theheat sink.

A power electronics converter may be characterized by a converterparameter φ, defined as a size in the z-direction of the heat sink gapdivided by a maximum electric field strength in the heat sink gap(referred to herein as the second maximum electric field strength):

$\begin{matrix}{\varphi = \frac{G_{2}}{E_{2}}} & (20)\end{matrix}$

-   -   where:    -   G₂=Size of heat sink gap in z-direction, and    -   E₂=Maximum electric field strength in heat sink gap.

Table 18 shows exemplary values for the converter parameter φ. Thevalues of φ are characteristically lower than the prior art andassociated with a high power density. Values of φ are quoted in units ofnm²/V (nano-m²/V, or ×10⁻⁹ m²/V).

TABLE 18 Peak Rated $\varphi = \frac{G_{2}}{E_{2}}$ Power (nm²/V) (kW)Example 1 Example 2 Example 3  50 0.05 1.00 15 100 0.05 1.00 15 150 0.051.00 15 200 0.05 1.00 15 400 0.05 1.00 15

Power electronics converters in accordance with the present disclosuremay have values of φ less than or equal to 20 nm²/V. The value of φ maybe greater or equal to 0.01 nm²/V, as lower values may be associatedwith, for example, greater risk of electrical breakdown. Values of φ maybe, for example, in the range 0.05 nm²/V to 5 nm²/V, as this may strikea good balance between power density and reliability.

As with the first maximum electric field strength E₁, the second maximumelectric field strength E₂ is a maximum homogenous electric fieldstrength, and thus, singularities and other highly localized maxima areexcluded. By way of example, in FIG. 2B, a second region 58 isillustrated for determining a second maximum electric field strength ofthe second electric field 56. The second region 58 is located inside theheat sink gap G2 and spaced away from any obstructions and edges insidethe heat sink gap G2, such as those associated with the prepackages 12.Where the maximum electric field strength is determined by modelling(e.g., finite element analysis (FEA)), the maximum electric fieldstrength may be determined at a location at least three mesh cells fromany singularity in the model.

Table 19 provides exemplary values for the size G₂ of the heat sink gapbetween the multi-layer planar carrier substrate 11 and the heat sink 15in the z-direction and the maximum electric field strength E₂ in theheat sink gap.

TABLE 19 Peak Rated G₂ (mm) E₂ (kV/mm) Power Example Example ExampleExample Example Example (kW) 1 2 3 1 2 3 50 0.5 1.5 3.0 0.2 1.5 10 1000.5 1.5 3.0 0.2 1.5 10 150 0.5 1.5 3.0 0.2 1.5 10 200 0.5 1.5 3.0 0.21.5 10 400 0.5 1.5 3.0 0.2 1.5 10

FIGS. 3A and 3B illustrate the smallest cuboidal volume that enclosesthe commutation cell of a power electronics converter. Once again, theexample of a one-phase, two-level converter is chosen for ease ofexplanation. Reference is made to Equation 1, which defines theinductance-volume parameter α in terms of the smallest cuboidal volumethat encloses the commutation cell.

FIG. 3A is the cross-sectional view also shown in FIG. 2A. The x- andz-directions are indicated. The box with dashed lines, labelled A_(x-z),is a cross-section in the x-z plane through the smallest volume thatencloses the commutation cell of the converter 10. The cross-sectionalarea A_(x-z) encloses both the power circuit (e.g., the powersemiconductors 121 that are formed in the prepackages 12, the capacitor14, the DC inputs, and the AC outputs and the connections therebetween)and the gate driver circuit 13 and its connections with the powersemiconductors 121. The area A_(x-z) may be calculated as the product ofthe size of the box measured in the x-direction (L_(x)) and the size ofthe box measured in the z-direction (L_(z)).

FIG. 3B is a plan view of converter of FIG. 3A. The x- and z-directionsare indicated. The box with dashed lines, labelled A_(x-y), is across-section in the x-y plane through the smallest cuboidal volume thatencloses the commutation cell of the converter 10. The cross-sectionalarea A_(x-y) encloses both the power circuit (e.g., the powersemiconductors 121, the capacitor, the DC inputs, and the AC outputs andthe connections therebetween) and the gate driver circuit 13 and itsconnections with the power semiconductors. The area A_(x-z) may becalculated as the product of the size of the box measured in thex-direction (L_(x)) and the size of the box measured in the y-direction(L_(y)).

The smallest cuboidal volume that encloses the commutation cell may becalculated as the product of the three dimensions, L_(x), L_(x), andL_(z).

All components of the commutation cell are visible in FIG. 3B eventhough some of the components illustrated in the cross-sectional view ofFIG. 3A are located on the underside of carrier substrate 11. This issolely for the purpose of illustration and clear explanation, and inpractice, not all components of the converter 10 of FIGS. 3A-3B would bevisible in a single plan view. The x-y position of the gate drivercircuit 13 has been moved in FIG. 3B for ease of illustration (e.g., soas not overlap with the x-y locations of the prepackages 12 as it doesin FIG. 3A). However, flexibility in the x-y location of commutationcell components is an advantageous feature of converters according tothe present disclosure; this is in part because changes in the x-ylocations do not significantly alter the parasitic inductance of theconnections between components.

While only two power semiconductor prepackages 12L and 12H are visiblein FIG. 3A, there are six prepackages 12L₁₋₃ and 12H₁₋₃ in FIG. 3B. Thisis to illustrate parallelization of the power semiconductor switchingelements 121. Specifically, FIG. 3B shows that the low side of thehalf-bridge includes three power semiconductor prepackages 12L₁, 12L₂,and 12L₃, including a corresponding three power semiconductor switchingelements connected together in parallel. Likewise, the high side of thehalf-bridge includes three power semiconductor prepackages, 12H₁, 12H₂,and 12H₃, including a corresponding three power semiconductor switchingelements connected in parallel. In FIG. 3A, where the y-direction isobscured, only one of three prepackages 12L, 12H of each of the low andhigh side are visible.

FIG. 4A illustrates a single power semiconductor prepackage 12 and showsdetail not visible in FIGS. 2A and 3A.

The power semiconductor switching element 121, which in this example isa MOSFET in the form of a semiconductor die or chip, is embedded insolid insulating material 122 (e.g., FR4). Electrical connections 124,125 i that may be filled holes, vias, or similar extend in thez-direction from the terminals of the semiconductor switching element121 to the electrical connection side 123 a of the prepackage 12, wherethe electrical connections 124, 125 i terminate to form a flat surface123 a. Although vertically extending connections 124, 125 i areillustrated, it should be appreciated the connections may have acomponent in the x-y plane too.

The MOSFET 121 has at least three terminals (e.g., the source, drain,and gate terminals). A first electrical connection 124 extends from thesource terminal to the flat electrical connection surface 123 a. In thisexample, the drain and gate terminals are electrically connected by anelectrically conductive metallization layer 125 ii on the underside ofthe MOSFET die 121. A second electrical connection 125 i extends fromthe conductive layer 125 ii to the flat electrical connection surface123 a. In other examples, the gate and drain terminals are not connectedand, for example, the connections 125 i, 125 ii correspond only to thedrain terminal, with the gate terminal served by a separate connectionfrom the gate terminal to the flat electrical connection surface 123 a.

The illustrated prepackage 12 further includes an optional electricalisolation layer (EIL) 126. The purpose of the EIL 126, which in thisexample is a layer of ceramic material, is to electrically isolate theMOSFET 121 and its terminals from the heat sink 15 that is arranged onthe underside of the prepackage 12 (see FIG. 4B). In other embodiments,the EIL 126 may be omitted from the prepackage 12, for example, if aseparate EIL is provided between the underside of the prepackage 12 andthe heat sink 15, or if the TIL 16 is able to provide sufficientelectrical isolation. Where present, the EIL 126 may have a thickness ofless than 5 mm. In one embodiment, the EIL 126 has a thickness of lessthan 1 mm to keep the converter compact while still achieving theisolation function. Lower thicknesses (e.g., less than 0.1 mm) may beused where, for example, an organic substrate such as IMS is used.

The illustrated prepackage 12 further includes an optional metal layer127 on the underside of the EIL 126. The metal layer 127 improvesthermal conduction between the prepackage 12 and TIL 16. The metal layer127 may also provide a suitable material interface between the undersideof the prepackage and TIL 16. For example, if the TIL 16 is a solderlayer, this may necessitate that the underside 123 b of the prepackage12 carries a material suitable for a solder connection. The metal layer127 may be omitted (e.g., if a TIL other than solder is used).

FIG. 4B shows the prepackage 12 of FIG. 4A sandwiched between amulti-layer carrier substrate 11 and a heat sink 15. FIG. 4B furtherillustrates how the electrical connections 124, 125 ii that extend fromthe terminals of the power semiconductor 121 to the electricalconnection surface 123 a may be connected to the conductive elements ofthe multi-layer carrier substrate 11 and thereby other components of thecommutation cell.

In the illustrated example, a planar surface 111 b of the multi-layercarrier substrate 11 includes one or more regions carrying an outerconductive layer 1121. These regions 1121 allow for soldered or sinteredconnections 113 to be formed to connect the substrate 11 to the distalends of the electrical connections 124, 125 i of the prepackage 12. Inthe illustrated example, metallization regions 1241 (e.g., conductivecontact pads such as solder pads) are also provided adjacent to thedistal ends of the electrical connections 124, 125 i to improve the easewith which soldered, sintered, or glued connections may be formed. Inother examples, these may be omitted, and the solder connections 113 maybe made directly on the exposed distal ends of the connections 124, 125ii.

With the terminals of the power semiconductor switching elements 121 nowconnected to the conductive regions 1121 of the carrier substrate 11,connections to the other components of the commutation cell are madethrough connection to the conductive regions 1121. These connections maybe formed by a combination of conductive layers 112 of the carriersubstrate 11 (see FIG. 2A) and electrical connections 114 that extend inthe z-direction from the conductive regions 1121 through the carriersubstrate. For example, referring to FIGS. 2A and 4B, the layer 1121that is soldered to the connection 124 that connects to the sourceterminal of the MOSFET 121 may be associated with one or more (e.g.,many) connections 114 a that extend in the z-direction to an internalconductive layer 112. The internal conductive layer 112 may then connectto a terminal of the DC link capacitor 14. As another example, the layer1121 that is soldered to the connection 125 i that connects to the gateterminal of the MOSFET 121 may be associated with one or moreconnections 114 b that extend in the z-direction all the way through thesubstrate 11 to terminals of the gate driver circuit 13.

Power electronics converters described herein may be characterized by aconverter parameter ρ, defined as follows:ρ=k×E _(Break)  (21)

-   -   where:    -   k=thermal conductivity of thermal isolation layer (TIL), and    -   E_(Break)=breakdown electric field strength of electrical        isolation layer (EIL).

Table 20 shows exemplary values for the converter parameter ρ.Converters described herein may have characteristically high values of ρthat may be associated with a combination of good heat removal from theprepackages and good resistance to electrical breakdown. Values of ρ arequoted in units of MVW/m²K (Mega-VW/m₂K, or ×10⁶ VW/m²K).

TABLE 20 ρ = k × E_(Break) Peak Rated (MVW/m²K) Power (kW) Example 1Example 2 Example 3 Example 4 50 10 70 4.5 × 10³ 22.5 × 10³ 100 10 704.5 × 10³ 22.5 × 10³ 150 10 70 4.5 × 10³ 22.5 × 10³ 200 10 70 4.5 × 10³22.5 × 10³ 400 10 70 4.5 × 10³ 22.5 × 10³

Power electronics converters in accordance with the present disclosuremay have values of ρ greater than or equal to 5 MVW/m²K, though valuesgreater than 20 MVW/m²K may be provided.

Table 21 shows exemplary values for the breakdown electric fieldstrength (which may also be referred to as the dielectric strength insome literature), E_(Break), of the electrical isolation layer (EIL).For exemplary values for thermal conductivity k of the thermal interfacelayer, see Table 15.

TABLE 21 E_(Break) Peak Rated (kV/mm) Power (kW) Example 1 Example 2Example 3 Example 4 50 10 20 50 250 100 10 20 50 250 150 10 20 50 250200 10 20 50 250 400 10 20 50 250

Higher values for E_(Break) (e.g., in Example 4) are for EILs includingorganic materials, whereas the lower values (e.g., Examples 1, 2, and 3)are EILs including inorganic materials.

As noted above, the EIL 126 is optional. In alternative embodiments, theEIL 126 is omitted and a TIL 16 with suitably electrically isolatingproperties is provided. Thus, the TIL 16 may provide electricalisolation between the prepackages and the heat sink, as well as a goodheat path between the prepackages and heat sink.

Such embodiments may be characterized by a TIL parameter λ, defined asthe thermal conductivity of the TIL divided by the electricalconductivity of the TIL:

$\begin{matrix}{\lambda = \frac{k}{P}} & (22)\end{matrix}$

-   -   where:    -   k=thermal conductivity of thermal isolation layer (TIL), and    -   P=electrical conductivity of the TIL.

Table 22 shows exemplary values of the parameter A. The values of λ arecharacteristically high. Values of λ are quoted in units of TW/SK(Tera-W/SK, or ×10¹² W/SK).

TABLE 22 Peak Rated $\lambda = {\frac{k}{P}\left( {{TW}/{SK}} \right)}$Power (kW) Example 1 Example 2 Example 3  50 1 350 90 × 10³ 100 1 350 90× 10³ 150 1 350 90 × 10³ 200 1 350 90 × 10³ 400 1 350 90 × 10³

Power electronics converters in accordance with the present disclosuremay have values of λ greater than or equal to 1 TW/SK, though valuesgreater than 100 TW/SK may be provided.

Table 23 shows exemplary values for the electrical conductivity P of thethermal interface layer.

TABLE 23 P (S/m) Peak Rated Example Example Example Power (kW) 1 2 3 501 × 10⁻¹⁵ 1 × 10⁻¹⁴ 1 × 10⁻¹³ 100 1 × 10⁻¹⁵ 1 × 10⁻¹⁴ 1 × 10⁻¹³ 150 1 ×10⁻¹⁵ 1 × 10⁻¹⁴ 1 × 10⁻¹³ 200 1 × 10⁻¹⁵ 1 × 10⁻¹⁴ 1 × 10⁻¹³ 400 1 ×10⁻¹⁵ 1 × 10⁻¹⁴ 1 × 10⁻¹³

FIG. 4C illustrates a further arrangement of a power electronicsconverter 10 in which at least a portion of the prepackage gap G1 isfilled with electrically insulating material 60 (e.g., a resin). Theelectrically insulating material 60 is arranged in and, optionally,around the prepackage gap G1. “Around” in this context provides that theelectrically insulating material 60 extends beyond the semiconductorprepackage 12 in the x-y direction. For example, the material 60 mayadditionally cover the metal layers 1121 extending from connections 114a to the connections 113.

Using the electrically insulating material 60, a creepage distance 62 aswell as an air gap distance 64 may be reduced and kept physically small.Thus, utilizing the electrically insulating material 60, small distancesbetween the components of the power electronics converter 10 (e.g.,between the metal layers 1121 and the connections 114 a) may be achievedwithout a significant risk of adverse electrical effects such assparking or creeping currents. This is particularly advantageous giventhe high voltages utilized in converters of the present disclosure,which may result in high potential differences between, for example, themetal layers 1121 and the prepackage surfaces 12 and heat sink 15.

The applied electrically insulating material 60 may include voids (e.g.,air) within its volume due to imperfections in the manufacturingprocess. A power electronics converter utilizing electrically insulatingmaterial in a prepackage gap may be characterized by a converterparameter σ, defined as an insulation fill factor divided by a maximumvoid size of the voids:

$\begin{matrix}{\sigma = \frac{F}{R_{\max}}} & (23)\end{matrix}$

-   -   where:    -   F=Insulation fill factor, and    -   R_(max)=Maximum void size in insulation.

In this equation, F is the insulation fill factor and R_(max) is themaximum void size of the plurality of voids. The insulation fill factoris defined as a cumulated volume of the plurality of voids (the ‘voidvolume’), subtracted from a volume of the electrically insulatingmaterial, divided by the volume of the electrically insulating material.This may be expressed as follows:

$\begin{matrix}{F = \frac{V_{IM} - V_{V}}{V_{IM}}} & (24)\end{matrix}$

-   -   where:    -   V_(IM)=Volume of electrically insulating material (including        voids), and    -   V_(V)=Cumulated volume of voids.

Hence, the converter parameter σ may also be expressed as:

$\begin{matrix}{\sigma = \frac{V_{IM} - V_{V}}{V_{V} \times R_{\max}}} & (25)\end{matrix}$

Table 24 shows exemplary values for the converter parameter σ, expressedin units of 1/mm.

TABLE 24 Peak Rated $\sigma = \frac{F}{R_{\max}}$ Power (1/mm) (kW)Example 1 Example 2 Example 3  50 18 100 1000 100 18 100 1000 150 18 1001000 200 18 100 1000 400 18 100 1000

Converters according to the present disclosure may have values of σgreater than or equal to 10/mm to provide good electrical insulationproperties. However, values greater than or equal to 50/mm may beprovided, especially at higher operating voltages.

Table 25 shows exemplary values for the insulation fill factor F and themaximum void size R_(max).

TABLE 25 Peak Rated F (%) R_(max) (μm) Power Example Example ExampleExample Example Example (kW) 1 2 3 1 2 3 50 90 99 99.99 1 10 50 100 9099 99.99 1 10 50 150 90 99 99.99 1 10 50 200 90 99 99.99 1 10 50 400 9099 99.99 1 10 50

Values of R_(max) may be determined through an equivalent-sphere methodin which measurements of the void size are made for a representativesample of the electrically insulating material, and a maximum void sizeis statistically estimated under the assumption the voids are sphericaland the measured sizes are diameters of spheres.

A power electronics converter utilizing electrically insulating materialin a prepackage gap may also be characterized by a converter parameterτ, defined as the product of the dielectric strength of the electricallyinsulating material and the maximum void size:τ=D×R _(max)  (26)

-   -   where:        -   D=Dielectric strength of electrically insulating material,            and        -   R_(max)=Maximum void size in insulation.

Table 26 shows exemplary values for the converter parameter τ, expressedin units of Volts.

TABLE 26 τ = D × R_(max) Peak Rated (V) Power (kW) Example 1 Example 2Example 3 50 1 200 10,000 100 1 200 10,000 150 1 200 10,000 200 1 20010,000 400 1 200 10,000

Converters according to the present disclosure may have values of τ lessthan or equal to 1,000 V to provide good electrical insulationproperties. Values less than or equal to 100 V may, however, beprovided, especially at higher operating voltages.

Table 27 shows exemplary values for the dielectric strength D of theelectrically insulating material.

TABLE 27 D Peak Rated (kV/mm) Power (kW) Example 1 Example 2 Example 350 1 20 200 100 1 20 200 150 1 20 200 200 1 20 200 400 1 20 200

FIG. 5A illustrates how low-side and high-side power semiconductorswitching elements 121L, 121H of the prepackages 12L, 12H may beelectrically connected using the multi-layer carrier substrate 11. Thex- and z-directions are indicated.

As in FIG. 4B, a planar surface 111 b of the substrate 11 has regions1121 i, 1121 ii, 1121 iii carrying an outer conductive layer. Theseregions 1121 i-iii facilitate connection to the terminals of the powersemiconductors 121L, 121H by soldering or sintering. The multi-layersubstrate 11 is further shown to include an internal conductive layer112 a that electrically connects to one of the outer layer regions 1121ii through a set of connections 114 a that extends in the z-directionthrough the substrate 11.

The source terminal (S) of the high-side power semiconductor switchingelement 121H is electrically connected to the high-side DC input (DC+)through a connection (e.g., a soldered, sintered, or glued connection)to the third outer layer region 1121 iii. The drain terminal (D) of thelow-side power semiconductor switching element 121L is electricallyconnected to the low-side DC input (DC−) through a connection to thefirst outer layer region 1121 i. The drain terminal (D) of the high-sidepower semiconductor switching element 121H and the source terminal (S)of the low-side power semiconductor switching element 121L areelectrically connected to each other and to the inner conductive layer112 of the substrate by connections to the second outer layer region1121 ii.

The internal layer 112 is shown to be thicker in the z-direction thanthe outer layer regions 1121 i-1121 iii. This increased thicknessreduces the resistance and thus increases the current carryingcapability of the inner conductive layer 112. This reflects the factthat, in this example, the inner conductive layer 112 carries a highcurrent, whereas the outer layer regions 1121 i-iii are used aselectrical contact and not paths for carrying current betweencomponents. The thin outer layer regions 1121 i-1121 iii may have athickness of less than 100 μm (e.g., 50 μm), whereas the thicker innerlayer 114 may have a thickness of greater than 100 μm (e.g., 100-400μm). A plurality (e.g., five, ten, or more) of vias may be used toconnect a thin outer contact region and the thick inner layer. Byincreasing the number of vias for one electrical path, the currentcarrying capability may be increased accordingly.

FIG. 5B illustrates how power semiconductor switching elements P1 and P2may be connected in parallel. P1 and P2 may, for example, be twolow-side power semiconductors of one phase of a two-level AC-DCconverter. The y- and z-directions are indicated for comparison withFIG. 5A.

As in FIG. 5A, a planar surface 111 b of the substrate 11 has regions1121 iv, 1121 v, 1121 vi carrying an outer conductive layer. Theseregions 1121 iv, 1121 v, 1121 vi facilitate connection to the terminalsof the power semiconductors P1 and P2 by, for example, soldering,sintering, or gluing. The multi-layer substrate 11 is further shown toinclude two internal conductive layers 112 b, 112 c. Each of theinternal layers 112 b, 112 c electrically connects to one of the outerlayer regions 1121 v, 1121 vi through corresponding sets of connections114 b, 114 c that extend in the z-direction through the substrate 11. Asin FIG. 5A, the internal layers 112 b, 112 c are thicker than the outerlayer regions 1121 iv-vi.

The source terminal (S) of the first power semiconductor switchingelement P1 is electrically connected to the AC side of the converter 10through a connection to the first outer layer region 1121 iv. The sourceterminal (S) of the second power semiconductor switching element P2 iselectrically connected to the AC side of the converter 10 through aconnection to the third outer layer region 1121 vi, which connects on tothe ticker inner conductive layer 112 c through the connection 114 c.The drain terminal (D) of the first power semiconductor switchingelement P1 and the drain terminal (D) of the second power semiconductorswitching element P2 are electrically connected to each other and to theinner conductive layer 112 b of the substrate 11 by soldered, sintered,or glued connections to the second outer layer region 1121 v.

The connection arrangements of FIGS. 5A-5B are merely examples, andconnections may be made in various different ways, with differentcombinations of outer layers, inner layers, and z-direction connections.The various internal layers 112 a-c are shown to be of the samethickness and at the same depth through the substrate 11. This is notnecessary and may not be the case. Internal layers may have differentthicknesses and be staggered in the z-direction and/or x-y direction.The outer layers 1121 i-vi and inner layers 112 a-c may be of the sameor different thicknesses. Suitable thicknesses will depend to someextent on the application requirements (e.g., power). It may be easierto fabricate thick internal layers 112 a-c through, for example, knownPCB manufacturing techniques than thick outer layers that may beprovided by, for example, deposition.

In each of the examples described above, the AC-DC converter 10 has onlya single phase. This, however, is only for ease of illustration andexplanation, and AC-DC converters according to the present disclosuremay have multiple phases. To this end, FIGS. 6A-6B illustrate how theconcepts described above may be extended to multiple phases. FIGS. 6A-6Billustrate a two-level, three-phase AC-DC converter in which thelow-side and high-side of each phase includes multiple (e.g., eight)parallel-connected power semiconductor switching elements.

FIG. 6A is a schematic cross-section of the converter 10. Thex-direction and z-direction are indicated. The converter 10 has threephases, designated U, V, and W. Each phase has its own set of powersemiconductor prepackages: the first phase U has prepackages 12U-L and12U-H; the second phase V has prepackages 12V-L and 12V-H; and the thirdphase W has prepackages 12W-L and 12W-H. The prepackages 12 of eachphase are surface mounted and electrically connected to a commonmulti-layer planar carrier substrate 11 that, as before, may be a PCB.The heat removal sides of the prepackages 12 face a heat sink 15, withthermal interface layers 16U, 16V, 16W between the undersides of theprepackages 12U-W and the heat sink 15.

For ease of illustration, the other components of the commutation cell(e.g., the gate driver circuit 13, the capacitor(s) 14, and theelectrical connections between the components) are not shown. Thesecomponents and their connections will be substantially as describedabove with reference to FIGS. 1-5 .

In this example, there is a common heat sink 15 that serves the entireconverter 10, but there may instead be a separate heat sink for eachphase U, V, W, similar to the arrangement shown in FIG. 2 for a singlephase. Further, in this example, the prepackages sit in a recess 151 inthe heat sink 15. The use of a recess may reduce the thickness of theconverter in the z-direction and may also provide a secondary thermalconduction path between the prepackages 12 and the heat sink 15 thatpasses through the substrate 11. Further, provided the recess issuitably sealed, a liquid or gaseous cooling medium may be flowedthrough the recess to directly cool the prepackages 12.

In the illustrated example, the heat sink 15 and substrate 11 aresecured and pressed together by fasteners 17. This is not essential, butthe use of fasteners to press the substrate 11 and prepackages 12 to theheat sink 15 may be provided for a better thermal interface to the heatsink 15.

The heat sink is shown to define barrier walls 152 that separate therecess 151 into three chambers (e.g., one for each phase) to provideisolation between the phases. This may be useful for fault mitigation,but in other examples, may be omitted. The barrier walls 152 may alsonot be integral with the heat sink 15, though integral barrier walls mayimprove the quality of the secondary thermal conduction path between theprepackages 12 and heat sink 15. Where the barrier walls 152 areomitted, a TIL 16 spanning the prepackages of multiple (e.g., all) ofthe phases U, V, W of the converter 10 may be used.

FIG. 6B is a schematic plan view of the converter 10 of FIG. 6A. Onlythe prepackages 12 are illustrated and, as with FIG. 3B, the prepackages12 are shown despite being located on the underside of the substrate andbeing obscured by the heat sink 15. With the y-direction no longerobscured, FIG. 6B shows that the low-side and high-side of each phase U,V, W includes eight power semiconductor switching elements each providedin a power semiconductor prepackage. For example, the eight prepackages12U-L₁ to 12U-L₈ are labelled.

Those skilled in the art will appreciate that the example of FIGS. 6A-Bmay be extended to any number of phases and to any number ofparallel-connected power semiconductor switching elements. The exampleof FIGS. 6A-6B may also be extended to DC-DC converter circuits, whichmay only include a single power semiconductor switching element or oneset of parallel-connected power semiconductor switching elements.

A power electronics converter may be formed of one or more ‘logicalswitches’ each including one or more parallel-connected powersemiconductor switching elements. In the case of a two-level AC-DCconverter, there are two logical switches per phase (e.g., one low-sideand one high-side logical switch). In the case of a DC-DC converter,there may be as few as one logical switch (see, e.g., FIG. 10A), thoughother DC-DC converters may include multiple logical switches (see, e.g.,FIG. 10B that has four logical switches; two per side of the transformer250′). Converters according to the present disclosure may include anynumber of power semiconductors (and thus prepackages) per logicalswitch, (e.g., three to ten prepackages per logical switch).

It is worth considering how a change to the number of phases affects thedefinition of the volume of the commutation cell and the parasiticinductance of the power circuit. Each phase forms part of thecommutation cell. Thus, the smallest cuboidal volume that enclose thecommutation cell will enclose every phase of the converter. However,each phase circuit is essentially independent from the other phasecircuits, with its switching and the conduction between its DC and ACsides being independent of the other phase circuits. Thus, a multi-phasepower circuit may, from the perspective of parasitic inductance, beconsidered equivalent to multiple independent one-phase power circuits,and the parasitic inductance of the power circuit is therefore equal tothe parasitic inductance of one of the phases. The parasitic inductanceof each phase will be the same (except for small unavoidable variationdue to, for example, component manufacturing tolerance and electricalcontact quality), so it does not matter which phase is selected. Inprinciple, it is possible to intentionally design a converter in whicheach phase has a different parasitic inductance, but this would beundesirable.

By way of specific examples, Table 28 includes specifications of twoexample converters in accordance with the present disclosure. Both aretwo-level, three-phase AC-DC converters, but it will be understood thisis not intended to be limiting.

TABLE 28 Example 1 (100 Example 1 (200 kW, 2-Level, 3- kW, 2-Level, 3-Phase AC-DC) Phase AC-DC) Substrate Type Rigid Multi-layer RigidMulti-layer PCB PCB Power Semiconductor SiC MOSFET SiC MOSFET TypePrepackage Type FR4 isolation FR4 isolation with integral with integralceramic EIL ceramic EIL Heat Sink Type Air-cooled Liquid-cooled aluminumaluminum TIL Type Indium-tin Indium-tin foil solder Peak Power Rating100 kW 200 kW MOSFET Source-Drain 1,200 V 1,400 V Blocking Voltage PeakRated Current 200 A 350 A Maximum Switching 50 kHz 50 kHz FrequencyMaximum Source-Drain 30 kV/μs 35 kV/μs Voltage Ramp Rate [dv/dt]Parasitic Inductance of 4 nH 2 nH Power Circuit [L] Total Power Circuit50 μF 100 μF Capacitance [C, @ 298K, 1,000V DC] Commutation Cell 300 cm³424 cm³ Volume [Smallest Cuboidal] Number of Prepackages 6 12 perLogical Switch Total Number of Pre- 36 72 packages Gap Between Substrate100 μm 120 μm and Prepackages Gap Between Substrate 1.5 mm 1.6 mm andHeat Sink TIL Thickness 200 μm 150 μm EIL Thickness 0.25 mm 0.25 mm TILThermal Con- 3.5 W/mK 2.5 W/mK ductivity Efficiency 99% 99% α = L × Vol1.2 pHm³ 0.85 pHm³ β = f_(max) × |dv/dt|_(max) 1.5 PV/s² 1.75 PV/s²$\gamma = \frac{C}{P \times f_{\max}}$ 10 fFs/W 10 fFs/W$\delta = \frac{V_{block}}{L \times C}$ 6 PV/FH 7 PV/FH$\varepsilon = \frac{f_{\max} \times {❘\frac{dv}{dt}❘}\max}{L \times C}$7.5 × 10²⁷ V/s⁴ 8.8 × 10²⁷ V/s⁴ $\theta = \frac{G_{1}}{E_{1}}$ 6.25pm²/V 6.25 pm²/V $\varphi = \frac{G_{2}}{E_{2}}$ 1 nm²/V 1 nm²/V$\sigma = \frac{F}{R_{\max}}$ 80/mm 100/mm τ = K × R_(max) 20 V 20 V ρ =k × E_(Break) 25 GV/m²K 25 GV/m²K $\eta = \frac{h}{G_{3}}$ 0.3 MW/m³K0.7 MW/m³K $\Omega = \frac{M}{K}$ 0.8 MNK/Wm 0.8 MNK/Wm

FIGS. 7A-7B and FIGS. 8A-8B illustrate a number of alternative ways inwhich the power semiconductor prepackages 12 may be arranged andconnected with respect to the multi-layer carrier substrate 11. In eachcase, the gate driver circuit 13 and capacitor(s) 14 are omitted, but itwill be understood these may be incorporated into the alternativearrangements substantially as described above.

FIG. 7A shows a converter arrangement 10 described above with referenceto FIGS. 2A, 2B, 3A, 3B, 4A-4C, 5A, 5B, 6A, and 6B and in Table 28. Eachpower semiconductor prepackage 12 is surface mounted on the underside ofthe multi-layer carrier substrate 11, with electrical connectionsbetween the two taking the form of soldered, sintered, or gluedconnections 113. The other side of the prepackages 12 faces a commonheat sink 15, with a TIL 16 providing the mechanical and thermalinterface between the prepackage 12 and the heat sink 15.

FIG. 7B shows an alternative converter arrangement 10′ in which theprepackages 12 and the heat sink 15 are mounted on opposite sides of thecarrier substrate 11. As before, the prepackages 12 are surface mountedto the substrate 11 using soldered or sintered connections 113. The heatsink 15 mechanically and thermally interfaces with the substrate 11 viaa TIL 16. The length and thermal resistance of the path of thermalconduction between the prepackages 12 and the heat sink 15 is increasedby mounting the prepackages 12 and heat sink 15 on opposite sides of thesubstrate 11. For this reason, the arrangement of FIG. 7A may beprovided instead of the arrangement of FIG. 7B. The addition ofintegrated thermally conductive elements 115 into the carrier substratemay, however, provide a useful and sufficient reduction in the thermalresistance. The thermally conductive elements may be, for example,copper vias.

FIGS. 8A and 8B show further alternative converter arrangements 10″,10′″ in which the prepackages 12 are embedded within the multi-layercarrier substrate 11. Embedding the prepackages 12 may offer a number ofadvantages. First, the embodiments eliminate some air gaps from theconverter that may otherwise be vulnerable to electrical breakdown dueto the high voltages and electric fields that may be used in convertersof the present disclosure. Second, it is possible to make the convertersomewhat more compact in the z-direction because of the elimination ofsome gaps. Third, there is a reduced need to form, for example, solderedor sintered connections between the prepackages and the conductiveelements of the substrates 11, which provides some reduction inmanufacturing complexity and eliminates a source of failure. Removingheat from the arrangements 10″, 10′″ of FIGS. 8A and 8B is, however,more challenging.

FIG. 8B differs from FIG. 8A in that the prepackages 12 are fullyembedded within the substrate 11 and thus entirely surrounded byinsulating material. In FIG. 8A, an underside of the prepackage 12 isexposed and flush with one of the planar surfaces of the substrate 11.Fully embedding the prepackages 12 in the substrate 11 results in afurther improvement in the electrical isolation of the powersemiconductors 121, but also further increases the thermal resistancebetween the prepackages 12 and heat sink 15. In FIG. 8B, thermallyconductive elements 115 are integrated into the substrate 11 to reducethe thermal resistance of the path of the thermal conduction.

In each of the examples described above, the power electronicsconverters 10, 10′, 10″, 10′″ have taken the form of a two-level AC-DCconverter. Those skilled in the art will appreciate that the conceptsdescribed may be equally applied to different types of power electronicsconverters, including alternative AC-DC converter topologies (includingmulti-level converter topologies) and DC-DC converters. FIGS. 9A-9C,10A, and 10B illustrate various AC-DC and DC-DC power electronicsconverter circuits that may be used in accordance with the embodimentsdescribed above.

FIG. 9A shows a three-phase electrical machine 140 connected to atwo-level, three-phase AC-DC converter 100. The electrical machine 140is configured as a motor, and the converter 100 is configured as aninverter; the arrangement of a generator and rectifier would be verysimilar.

In this example, one end of each phase winding 140 u-w of the motor 140is connected at a common point (e.g., the ‘star’ or ‘Y’ point), though,for example, a Delta connection arrangement of the windings 140 u-w mayalso be used. The other end of each phase winding is connected to acorresponding phase leg 110 u-w of the inverter 100 at a phaseconnection point. Each phase leg 110 u-w is further connected to highand low DC inputs DC-H, DC-L that may, for example, connect to a DC bussuch as the DC bus 330 of FIG. 12A. Each phase leg includes high- andlow-side transistors 112H, 112L and associated parallel diodes 112H-d,112L-d connected between the high- and low-side DC inputs DC-H, DC-L.

The inverter 100 further includes a smoothing DC-link capacitor 114 thatis connected between the high and low DC inputs DC-H, DC-L and a gatedriver circuit 113 that is connected to and configured to supplyswitching signals to the gate terminals of the transistors 112H, 112L.The gate driver circuit 113 may receive low-power control signals from acontroller (not illustrated) and amplifies the low-power signals tosupply the gate terminals with switching signals suitable forcontrolling the on/off state of the transistors 112H, 112L of the phaselegs 110 u-w.

In this example, the transistors 112L, 112H are MOSFETs (e.g., SiliconCarbide (SiC) MOSFETs). Thus, as will be appreciated by those skilled inthe art, the parallel diodes 112H-d, 112L-d associated with the MOSFETs133, 134 may not be discrete components but rather the so-called ‘bodydiodes’ of the MOSFETs (e.g., the inherent diode characters of theMOSFETs). In other examples, the diodes 112H-d,112L-d may be discretecomponents separate from the transistors 112H, 112L.

In use, the inverter 100 receives DC electrical power via the DCconnections DC-H, DC-L. The gate terminals of the transistors 112H, 112Lreceive switching signals from the gate driver circuit 113. As will beunderstood by those skilled in the art, the switching signals switch thetransistors 112L, 112H of each phase leg 110 u-w between conductive andnon-conductive (e.g., cony and ‘off’) states, commutating current fromthe upper and lower branches of each phase leg 110 u-w to the respectivephase winding 140 u-w of the motor 140. Timings and durations of theswitching are controlled so that AC electrical power is supplied to thephase windings 140 u-w of the motor 140 via the AC phase connectionpoints.

FIG. 9B illustrates another inverter circuit 100′. In this example, afour-phase electrical motor 140′ is supplied with electrical power froma two-level DC supply via an AC-DC power electronics converter 100′including four independent H-bridge circuits. For clarity, FIG. 10B onlyshows two of the four H-bridge circuits 110 u′, 110 v′, connected with acorresponding two of the four phases 140 u′, 140 v′ of the motor 140′.

Each H-bridge circuit (e.g., H-bridge circuit 110 u′) includes fourtransistors 112L-1′, 112H-1′, 112L-2′, 112H-2′ and associated paralleldiodes connected in an H-bridge configuration between the high and lowDC connections DC-H, DC-L and one of the phase windings 140 u′ of themotor 140′. A DC-link capacitor 113 is also connected between the DCconnections DC-H, DC-L. During operation, the DC connections supply DCelectrical power to the H-bridge circuit 110 u′, and the gate drivercircuit 113 supplies switching signals to the gate terminals of thetransistors 112L-1′, 112H-1′, 112L-2′, 112H-2′. The switching of thetransistors between their conductive and non-conductive states affectsinversion of the DC power to AC power for supply to the phase windings140 u′ of the motor 140′.

FIG. 9C shows an inverter 100″ circuit with a DC-side filter that ismore complex than the single DC-link capacitor of the converters 100,100′ of FIGS. 9A-B. The inverter 100″ is, like the inverter 100 of FIG.9A, a two-level three phase inverter. However, the inverter 100″includes a DC filter having three capacitors 114-1, 114-2, 114-3 andfour inductors 119-1, 119-2, 119-3, 119-4 connected between the DCconnections DC-L, DC-H and the converter phase legs 110 u-w. Anysuitable DC-side filter circuit may be used in accordance with theembodiments described herein.

FIG. 10A illustrates a DC-DC converter 200 of the boost type. Buck typeand buck-boost type DC-DC converters are also known.

The DC-DC converter 200 is connected to a battery 220 at one of itssides and on the other of its sides is connected to, for example, a DCpower channel such as the DC power channel 330 of FIG. 11A or the DCpower channel 530 of FIG. 14A.

The DC-DC converter 200 includes a transistor 212, a gate terminal (g)of which is connected to a gate driver circuit 213. As in the previousexamples, the transistor 212 is a MOSFET and the parallel diodeassociated with the transistor 212 may be an additional discrete diodeof the inherent body diode of the MOSFET. The converter circuit 200further includes a diode 218, a smoothing capacitor 214 (which may bereferred to as the input capacitor in the context of a DC-DC converter),and an inductor 219.

In use, the DC-DC converter circuit 200 receives DC power, either fromthe terminals of the battery 220 or from the DC connections DC-L, DC-H.The gate driver circuit 213 supplies the gate terminal (g) of thetransistor with switching signals to control the on/off state of thetransistor 212 to affect the desired voltage increase or decrease.

FIG. 10B illustrates another another type of DC-DC converter 200′. TheDC-DC converter 200′ is of the DC-AC-DC type and includes back-to-backinverter 210′ and rectifier 230′ stages with an intermediate transformer250′. In this example, each of the inverter 210′ and the rectifier 230′are of the H-bridge type, though other inverter and rectifier circuitsmay be used. Each have associated DC-side filters 214′, 234′.

In use, the DC-DC converter 200′ receives DC current from a DC currentsource (e.g., the energy storage system 230 or via the DC connectionsDC-H, DC-L). The gate terminals of the transistors of the H-bridgecircuits 210′, 230′ receive switching signals from the gate drivercircuit 213, which affects inversion of DC and rectification of AC tosupply current to or receive current from the windings of thetransformer 250′. The DC output by one of the first and second H-bridgecircuits 210′, 230′ is supplied to either the battery 220 or a DCnetwork via the DC connections DC-L, DC-H.

FIGS. 11A, 11B, 12A, 12B, 13, 14A, and 14B illustrate exemplary aircraftpower and propulsion systems 300, 400, 500 including power electronicsconverters. The power electronics converters may be of the efficient,power-dense types described herein.

FIG. 11A is a schematic illustration of a purely electric aircraftpropulsion system 300. The system 300 includes an electrical energystorage unit 320 (e.g., a high-voltage battery pack) that supplies a DCpower channel 330 with DC electrical power. The DC power channel 330supplies electrical power to electrical loads including an electricmotor 340 that drives rotation of a propulsor 350 (e.g., a propeller orducted fan). An inverter 310 converts the DC power from the DC powerchannel 330 to AC power for supply to the windings of the motor 340. Insome embodiments, the inverter 310 may be integrated with (e.g., share acommon housing structure with) the motor 340, and/or the motor 340 maybe integrated with the propulsor 350.

Although not illustrated, the electrical propulsion system 300 mayoptionally include a DC-DC converter connected between the terminals ofthe battery pack 320 and the DC power channel 330 to regulate thevoltage on the DC power channel. For example, the terminal voltage ofthe battery pack 320 will tend to drop (e.g., by a factor of up to two)as the battery pack 320 discharges from a maximum charge level to alower charge level. By way of an example, the voltage may drop from amaximum voltage level of 900 V to 450 V over the course of itsdischarge. A DC-DC converter may therefore be used to boost the terminalvoltage to maintain a constant voltage on the DC power channel 330.Other arrangements may omit the DC-DC converter and compensate for thevoltage drop and the associated power drop by increasing the currentdelivered to the loads (e.g., the motor 340).

The electrical power and propulsion system 300 illustrated in FIG. 11Aincludes only a single power channel (or power ‘lane’) and a singlepropulsor. In practice, a power and propulsion system may includemultiple channels and be of a more complex configuration. To illustrate,FIG. 12A shows an exemplary electrical vertical take-off and landing(eVTOL) aircraft 360 that has a distributed propulsion system that inthis case includes six propulsors 350 a-f. In this arrangement, eachpropulsor 350 a-f may be associated with its own power channel (e.g.,six power channels of the type shown in FIG. 11A). In another example,one or more of the propulsors 350 a-f may share a power channel so thatthere are fewer than six power channels.

Still referring to FIG. 12A, four of the propulsors 350 a-d are coupledto the wings 361 of the aircraft 360, while the remaining two propulsors350 e-f are coupled to rear flight surfaces 362. The wings 361 and rearflight surfaces 362 tilt between a VTOL configuration (shown in FIG.12A), in which the propulsors 350 a-f provide thrust for lift, and aforward flight configuration (shown in FIG. 12B), in which at least someof the propulsors (e.g., the rear propulsors 350 e-f in FIG. 12B)provide forward thrust. Other numbers of propulsors (e.g., four oreight) are possible, as are other eVTOL configurations (e.g.,multicopter designs and variations of the illustrated tilt rotor designare also known).

FIG. 11B illustrates how six electrically powered propulsors (e.g., thesix propulsors 350 a-f of FIGS. 12A-B) may be arranged within anelectrical power system 300. The power system 300 includes threeindependent power system channels 300 a, 300 b, 300 c, each of which isassociated with two of the six propulsors 350 a-f. In this specificexample, each of the first two power channels 300 a, 300 b is associatedwith one front propulsor 350 b, 350 c and one rear propulsor 350 e, 350f, while the third channel 300 c is associated with one front-leftpropulsor 350 a and one front-right propulsor 350 d. Each propulsor 350a-f is associated with a respective motor 340 a-f and a respectiveinverter 340 a-f. Other power system configurations are possible andwill occur to those skilled in the art. For example, the system 300 mayfeature connections between power channels 300 a-c and the use ofmulti-redundant motor winding arrangements to allow power sharingbetween some or all the power channels 300 a-c and to increasefault-tolerance and power availability.

The use of a distributed propulsion system such as that illustrated inFIG. 11B and FIGS. 12A-B may be highly desirable in terms of improvedflight characteristics, reduced aerodynamic noise, and reduced drag. Theuse of the distributed propulsion system, however, also may increase thenumber of power electronics converters present in the electrical powersystem. For example, applying the system of FIG. 11B, the platform mayinclude nine independent power electronics converters (e.g., sixinverters and three DC-DC converters), which may add considerable weightto the platform and add losses to the power system. Thus, for platformsof this type, the improvements in the efficiency and power-density ofthe converters disclosed herein may be particularly advantageous.

FIG. 13 shows an arrangement of an engine 400 for an aircraft. Theengine 400 is of turbofan configuration, and thus includes a ducted fan401 that receives intake air A and generates two pressurised airflows: abypass flow B that passes axially through a bypass duct 402 and a coreflow C that enters a core gas turbine.

The core gas turbine includes, in axial flow series, a low-pressurecompressor 403, a high-pressure compressor 404, a combustor 405, ahigh-pressure turbine 406, and a low-pressure turbine 407.

In operation, the core flow C is compressed by the low-pressurecompressor 403 and is then directed into the high-pressure compressor404 where further compression takes place. The compressed air exhaustedfrom the high-pressure compressor 404 is directed into the combustor 405where the compressed air is mixed with fuel and the mixture iscombusted. The resultant hot combustion products then expand through,and thereby drive, the high-pressure turbine 406 and in turn thelow-pressure turbine 407 before being exhausted to provide a smallproportion of the overall thrust.

The high-pressure turbine 406 drives the high-pressure compressor 404via an interconnecting shaft. The low-pressure turbine 407 drives thelow-pressure compressor 403 via another interconnecting shaft. Together,the high-pressure compressor 404, high-pressure turbine 406, andassociated interconnecting shaft form part of a high-pressure spool ofthe engine 400. Similarly, the low-pressure compressor 403, low-pressureturbine 407, and associated interconnecting shaft form part of alow-pressure spool of the engine 400. Such nomenclature will be familiarto those skilled in the art. Those skilled in the art will alsoappreciate that while the illustrated engine has two spools, other gasturbine engines have a different number of spools (e.g., three spools).

The fan 401 is driven by the low-pressure turbine 407 via a reductiongearbox in the form of a planetary-configuration epicyclic gearbox 408.Thus, in this configuration, the low-pressure turbine 407 is connectedwith a sun gear of the gearbox 408. The sun gear is meshed with aplurality of planet gears located in a rotating carrier. The pluralityof planet gears is meshed with a static ring gear. The rotating carrierdrives the fan 401 via a fan shaft 410. In alternative embodiments, astar-configuration epicyclic gearbox (in which the planet carrier isstatic, and the ring gear rotates and provides the output) may be usedinstead, and the gearbox 408 may be omitted entirely so that the fan 401is driven directly by the low-pressure turbine 407.

It is increasingly desirable to facilitate a greater degree ofelectrical functionality on the airframe and on the engine. To this end,the engine 400 of FIG. 13 includes one or more rotary electricalmachines (e.g., capable of operating both as a motor and as agenerator). The number and arrangement of the rotary electrical machineswill depend to some extent on the desired functionality. Someembodiments of the engine 400 include a single rotary electrical machine420 driven by the high-pressure spool (e.g., by a core-mounted accessorydrive 421 of conventional configuration). Such a configurationfacilitates the generation of electrical power for the engine and theaircraft and the driving of the high-pressure spool to facilitatestarting of the engine in place of an air turbine starter. Otherembodiments, including the one shown in FIG. 13 , include both a firstrotary electrical machine 420 coupled with the high-pressure spool and asecond rotary electrical machine 430 coupled with the low-pressurespool. In addition to generating electrical power and the starting theengine 400, having both first and second rotary machines 420, 430,connected by power electronics, may facilitate the transfer ofmechanical power between the high and lower pressure spools to improveoperability, fuel consumption, etc.

As mentioned above, in FIG. 13 , the first rotary electrical machine 420is driven by the high-pressure spool by a core-mounted accessory drive421 of conventional configuration. In alternative embodiments, the firstelectrical machine 420 may be mounted coaxially with the turbomachineryin the engine 400. For example, the first electrical machine 420 may bemounted axially in line with the duct between the low- and high-pressurecompressors 403 and 403. In FIG. 13 , the second electrical machine 430is mounted in the tail cone 409 of the engine 400 coaxially with theturbomachinery and is coupled to the low-pressure turbine 407. Inalternative embodiments, the second rotary electrical machine 430 may belocated axially in line with low-pressure compressor 403, which mayadopt a bladed disc or bladed drum configuration to provide space forthe second rotary electrical machine 430. It will be appreciated bythose skilled in the art that any other suitable location for the firstand, if present, second electrical machines may be adopted.

The first and second electrical machines 420, 430 are connected withpower electronics. Extraction of power from or application of power tothe electrical machines is performed by power electronics converters440. In the present embodiment, the power electronics converters 440 aremounted on the fan case 411 of the engine 400, but it will beappreciated that the power electronics converters 440 may be mountedelsewhere such as on the core of the gas turbine, or in the vehicle towhich the engine 400 is attached, for example.

Control of the power electronics converters 440 and of the first andsecond electrical machines 420 and 430 is in the present exampleperformed by an engine electronic controller (EEC) 450. In the presentembodiment, the EEC 450 is a full-authority digital engine controller(FADEC), the configuration of which will be known and understood bythose skilled in the art. The EEC 450 therefore controls all aspects ofthe engine 400 (e.g., both of the core gas turbine and the first andsecond electrical machines 420 and 430). In this way, the EEC 450 mayholistically respond to both thrust demand and electrical power demand.

The one or more rotary electrical machines 420, 430 and the powerelectronics converters 440 may be configured to output to or receiveelectric power from one, two, or more DC busses or power channels. TheDC power channels allow for the distribution of electrical power toother engine electrical loads and to electrical loads on the airframe.

Those skilled in the art will appreciate that the gas turbine engine 400described above may be regarded as a ‘more electric’ gas turbine enginebecause of the increased role of the electrical machines 420, 430compared with those of conventional gas turbines.

Turning now to FIG. 14A, this illustrates an exemplary power andpropulsion system 500 of a hybrid electric aircraft. The system 500includes a generator set 501 including an engine 560 and electricalgenerator 540 a, and a battery pack 520. Both the generator set 501 andthe battery pack 520 are used as energy sources to power a motor-drivenpropulsor 502, an example of which is shown in FIG. 14B.

The illustrated propulsion system 500 further includes a rectifier 510a, a DC distribution bus 530, an inverter 510 b, and a DC-DC converter510 c. It will be appreciated that while one generator set 501 and onepropulsor 502 are illustrated in this example, a propulsion system 500may include more than one generator set 501 and/or one or more propulsor502.

A shaft or spool of the engine 560 is coupled to and drives the rotationof a shaft of the generator 540 a, which thereby produces alternatingcurrent. The rectifier 510 a, which faces the generator 540 a, convertsthe alternating current into direct current that is fed to variouselectrical systems and loads via the DC distribution bus 530. Theseelectrical systems include non-propulsive loads (not shown in FIG. 14A)and the motor 540 b that drives the fan 550 of the propulsor 502 via theinverter 510 b.

The battery pack 520, which may be made up of a number of batterymodules connected in series and/or parallel, is connected to the DCdistribution bus 530 via the DC-DC converter 510 c. The DC-DC converter510 c converts between a terminal voltage of the battery pack 520 and avoltage of the DC distribution bus 530. In this way, the battery pack520 may replace or supplement the power provided by the generator set501 (e.g., by discharging and thereby feeding the DC distribution bus530) or may be charged using the power from the generator set 501 (e.g.,by being fed by the DC distribution bus 530).

Referring to FIG. 14B, in this example, the propulsor 502 takes the formof a ducted fan incorporating an electrical machine 540 b. The fan 550is enclosed within a fan duct 551 defined within a nacelle 552 and ismounted to a core nacelle 553. The fan 550 is driven by the electricalmachine 540 b via a drive shaft 554, both of which may also be thoughtof as components of the propulsor 502. In this embodiment, a gearbox 555is provided between the electrical machine 540 b and the drive shaft554.

The electrical machine 540 b is supplied with electric power from apower source (e.g., the generator set 501 and/or the battery 520 via theDC bus 530). The electrical machine 540 b of the propulsor 502, andindeed the electrical machine 540 a of the generator set 501, may be ofany suitable type (e.g., of the permanent magnet synchronous type).

The inverter 510 b may be integrated with (e.g., share a common housingstructure with) the electrical machine 540 b and thus form a part of thepropulsor 502. Likewise, the rectifier 510 a may be integrated with(e.g., share a common housing structure with) the electrical machine 540a. The DC-DC converter 510 c may itself be integrated with the energystorage system 520.

Those skilled in the art will recognize the propulsion system 500 ofFIGS. 14A-B to be of the series hybrid type. Other hybrid electricpropulsion systems are of the parallel type, while still others are ofthe turboelectric type or have features of more than one type. Theconfiguration of the more electric engine 400 of FIG. 13 may beconsidered similar to a parallel hybrid system, with the maindistinction being the roles of the electrical machines. For example, theelectrical machines of a more electric engine may only be used in motormode to start the engine and to improve engine operability, whereas theelectric machines of a parallel hybrid propulsion system are used tomotor the spools to meaningfully add to the amount of propulsive thrustproduced by the turbomachinery.

Those skilled in the art will also appreciate that the hybridarchitecture illustrated in FIG. 14A is only one example, and otherarchitectures are known and will occur to those skilled in the art.

Various examples have been described, each of which feature variouscombinations of features. It will be appreciated by those skilled in theart that, except where clearly mutually exclusive, any of the featuresmay be employed separately or in combination with any other features andthe disclosure extends to and includes all combinations andsub-combinations of one or more features described herein.

While the embodiments have been described with reference to an aircraft,and to turbofan engines, the principles of the described electricalsystems may be applied to other installations (e.g., to aircraft withturboprop and open rotor engines, to marine environments such as on anaval vessel, and to other transport applications including trains).

The invention claimed is:
 1. A power electronics converter comprising: aconverter commutation cell comprising a power circuit and a gate drivercircuit, the power circuit comprising at least one power semiconductorswitching element and at least one capacitor, wherein each powersemiconductor switching element of the at least one power semiconductorswitching element is comprised in a power semiconductor prepackage,wherein the power semiconductor prepackage comprises one or more powersemiconductor switching elements embedded in a solid insulatingmaterial, each power semiconductor switching element of the one or morepower semiconductor switching elements having at least three terminalsincluding a gate terminal, wherein the gate driver circuit iselectrically connected to each power semiconductor switching element ofthe at least one power semiconductor switching element, wherein the gatedriver circuit is configured to provide switching signals to the gateterminal of each power semiconductor switching element of the at leastone power semiconductor switching element, and wherein a peak ratedpower output of the power electronics converter is greater than 25 kWand a value of a converter parameter α is less than or equal to 5 pHm³,wherein the converter parameter a is a product of a smallest cuboidalvolume that encloses the converter commutation cell and a parasiticinductance of the power circuit of the converter commutation cell. 2.The power electronics converter of claim 1, wherein the value of theconverter parameter σ is greater than or equal to 0.3 pHm³.
 3. The powerelectronics converter of claim 1, wherein the value of the converterparameter α is less than or equal to 4 pHm³.
 4. The power electronicsconverter of claim 1, wherein the value of the converter parameter αsatisfies 0.4 pHm³≤α≤3.5 pHm³.
 5. The power electronics converter ofclaim 1, wherein the value of the converter parameter α satisfies 0.5pHm³≤α≤2.5 pHm³.
 6. The power electronics converter of claim 1, whereina product of the parasitic inductance of the power circuit of theconverter commutation cell and the peak rated power output is in a range0.05 mHW to 1.5 mHW.
 7. The power electronics converter of claim 1,wherein the value of the converter parameter α divided by the peak ratedpower output of the power electronics converter is in a range 1.5 aHm³/Wto 100 aHm³/W.
 8. The power electronics converter of claim 1, wherein atotal rated capacitance of the power circuit of the convertercommutation cell divided by the peak rated power output is less than orequal to 5 nF/W.
 9. The power electronics converter of claim 1, furthercomprising: a multi-layer planar carrier substrate defining an x-ydirection parallel to a planar surface of the multi-layer planar carriersubstrate and a z-direction perpendicular to the x-y direction, whereinthe multi-layer planar carrier substrate comprises a plurality ofelectrically conductive layers extending in the x-y direction and atleast one electrical connection extending in the z-direction, whereineach power semiconductor prepackage further comprises an electricalconnection from at least one terminal of the one or more powersemiconductor switching elements to an electrical connection side of therespective power semiconductor prepackage, wherein the electricalconnection from the at least one terminal extends in the z-directionthrough the solid insulating material, and wherein at least one terminalof each power semiconductor switching element of the one or more powersemiconductor switching elements is connected to at least oneelectrically conductive layer of the plurality of electricallyconductive layers of the multi-layer planar carrier substrate at theelectrical connection side of the respective power semiconductorprepackage.
 10. The power electronics converter of claim 9, wherein, foreach power semiconductor prepackage, the electrical connection side ofthe respective power semiconductor prepackage has a flat surface and therespective power semiconductor prepackage is surface mounted at theelectrical connection side of the respective power semiconductorprepackage to the planar surface of the multi-layer planar carriersubstrate.
 11. The power electronics converter of claim 10, wherein, foreach power semiconductor prepackage, each electrical connectionextending from at least one terminal of a power semiconductor switchingelement through the solid insulating material terminates at the flatsurface of the respective power semiconductor prepackage.
 12. The powerelectronics converter of claim 11, wherein each power semiconductorprepackage is surface mounted to the planar surface of the multi-layerplanar carrier substrate, and wherein, for each power semiconductorprepackage, a terminal electrical connection of the respective powersemiconductor prepackage is soldered, sintered, or glued to a respectiveelectrical connection of the multi-layer planar carrier substrate. 13.The power electronics converter of claim 12, wherein each soldered,sintered, or glued connection spaces apart a respective powersemiconductor prepackage from the planar surface of the multi-layerplanar carrier substrate to provide a respective gap, and wherein a sizeof the respective gap, as measured in the z-direction, is less than orequal to 300 μm.
 14. The power electronics converter of claim 9, whereinthe multi-layer planar carrier substrate comprises a rigid printedcircuit board (PCB), a flexible PCB, or a ceramic carrier substrate. 15.The power electronics converter of claim 1, wherein the powerelectronics converter is an AC-DC converter.
 16. The power electronicsconverter of claim 1, wherein the power electronics converter is a DC-DCconverter.
 17. An electrical propulsion unit (EPU) for an aircraft, theEPU comprising: an electric motor; and an AC-DC power electronicsconverter configured as an inverter and arranged to supply current to awinding of the electric motor, wherein the AC-DC power electronicsconverter comprises: a converter commutation cell comprising a powercircuit and a gate driver circuit, the power circuit comprising at leastone power semiconductor switching element and at least one capacitor,wherein each power semiconductor switching element of the at least onepower semiconductor switching element is comprised in a powersemiconductor prepackage, wherein the power semiconductor prepackagecomprises one or more power semiconductor switching elements embedded ina solid insulating material, each power semiconductor switching elementof the one or more power semiconductor switching elements having atleast three terminals including a gate terminal, wherein the gate drivercircuit is electrically connected to each power semiconductor switchingelement of the at least one power semiconductor switching element,wherein the gate driver circuit is configured to provide switchingsignals to the gate terminal of each power semiconductor switchingelement of the at least one power semiconductor switching element, andwherein a peak rated power output of the AC-DC power electronicsconverter is greater than 25 kW and a value of a converter parameter αis less than or equal to 5 pHm³, wherein the converter parameter α is aproduct of a smallest cuboidal volume that encloses the convertercommutation cell and a parasitic inductance of the power circuit of theconverter commutation cell.
 18. A gas turbine engine comprising: aspool; an electrical machine having a rotor mechanically coupled to thespool; and an AC-DC power electronics converter configured to supplycurrent to or receive current from a winding of the electric machine,wherein the AC-DC power electronics converter comprises: a convertercommutation cell comprising a power circuit and a gate driver circuit,the power circuit comprising at least one power semiconductor switchingelement and at least one capacitor, wherein each power semiconductorswitching element of the at least one power semiconductor switchingelement is comprised in a power semiconductor prepackage, wherein thepower semiconductor prepackage comprises one or more power semiconductorswitching elements embedded in a solid insulating material, each powersemiconductor switching element of the one or more power semiconductorswitching elements having at least three terminals including a gateterminal, wherein the gate driver circuit is electrically connected toeach power semiconductor switching element of the at least one powersemiconductor switching element, wherein the gate driver circuit isconfigured to provide switching signals to the gate terminal of eachpower semiconductor switching element of the at least one powersemiconductor switching element, and wherein a peak rated power outputof the AC-DC power electronics converter is greater than 25 kW and avalue of a converter parameter α is less than or equal to 5 pHm³,wherein the converter parameter α is a product of a smallest cuboidalvolume that encloses the converter commutation cell and a parasiticinductance of the power circuit of the converter commutation cell.